Training 1

Institut f¨ ur Integrierte Systeme Integrated Systems Laboratory

Department of Information Technology and Electrical Engineering

VLSI II: Entwurf von hochintegrierten Schaltungen 227-0147-00

Training 1

SoC Encounter for Designers II Prof. Dr. H. Kaeslin Dr. N. Felber

SVN Rev.: Last Changed:

1016 2013-10-15

Reminder: With the execution of this training you declare that you understand and accept the regulations about using CAE/CAD software installations at the ETH Zurich. These regulations can be read anytime at http://dz.ee.ethz.ch/regulations/index.en.html.

1 Overview Unlike other exercises in the VLSI lectures, the back-end design flow requires you to learn how to use a commercial Electronic Design Automation (EDA) tool, in our case C ADENCE S O C E NCOUNTER from Cadence Design Systems. These exercises are therefore called ’Trainings’ and will teach you the basics of C ADENCE S O C E NCOUNTER so that you can use it for your semester projects. There will be three trainings: • Training 1 Floorplanning, placement, clock tree synthesis, optimization, routing and timing analysis with C ADENCE S O C E NCOUNTER . • Training 2 Determining power consumption, IR drop analysis. • Training 3 Tape-out preparation, performing Design Rule Check (DRC) and Layout Versus Schematic (LVS) on your final database. Students who plan to work on an ASIC semester project should make sure to visit all three trainings.

1.1 About the Style We will try to use a number of different styles to identify different types of actions. These are summarized below: Student Task: Parts of the text that have a gray background, like the current paragraph, indicate steps required to complete the exercise. Actions that require you to select a specific menu fill be shown like the following: menu→sub-menu→sub-sub-menu Whenever there is an option or a tab that can be found in the current view/menu we will use a BUTTON to indicate such an option. Throughout the exercise you will be asked to enter certain commands using the commandline1 . The following is an example of the linux command line. sh > command to be entered on the linux command line Whereas some of the commands will be entered on the command line of the C ADENCE S O C E N COUNTER tool such as: enc > this command is an encounter command

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There are many reasons for using a commandline. Some functionality can not not be accessed through GUI commands, and in some cases, using the commandline will be much faster. Most importantly, things you enter on the commandline can be converted into a script and executed repeatedly

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2 Introduction In this training we will start with a structural Verilog design netlist (from synthesis) and create step by step a physical layout that can be manufactured. To keep runtimes reasonably low, we will use an example design with a (slightly) lower complexity than most student design projects.

2.1 Example Design The example design is based on the FIR filter that we have been using in the past exercises. The filter has been changed to include several pipelined filter stages as shown in the block diagram below2 .

ResetxRBI

DataInxDI

RamWDxD

DataInReqxSI

RamTestxTI

RamAddrxD

SY180_2048X16X1CM8 r256x72tb300xo

DataInAckxSO

RamRDxD

LUT 16

LUT

16

32

16

LUT

16

32

16

LUT

16

32

16

LUT

16

32

16

16

32

ScanEnxTI

DataOutAckxSI 48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

48

filter_stage1

filter_stage2

filter_stage3

filter_stage4

DataOutReqxSO

48

48

DataOutxDO

48

’0’ ClkxCI filter_stage8

filter fiter_top filter_chip

Each filter stage contains a large multiplier, a look-up table and an accumulator. Note that the input of the first stage is tied to constants and therefore greatly simplified. The following is a short description of all pins of the circuit:

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The filter is basically useless and has only been engineered as an example circuit suitable for the exercise.

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Pin Descriptions Name

Bits

Dir

Description

ClkxCI

1

In

Clock input

ResetxRBI

1

In

Reset input, active low signal, 0: Reset

ScanEnxTI

1

In

Scan Enable for testing, 1: Scan

RamTestxTI

1

In

Ram bypass control, 1: Test (RAM bypassed)

DataInxDI

16

In

16-bit data input

DataInReqxSI

1

In

Request signal for data input

DataInAckxSO

1

Out

Acknowledge signal for data input

DataOutxDO

16

Out

16-bit data output

DataOutReqxSO

1

Out

Request signal for data output

DataOutAckxSI

1

In

Acknowledge signal for data output

3 Getting Started You will need a terminal program to type in commands throughout this exercise. In the computers in the ETZ D61.2 you can get a terminal by accessing the menu on the top left corner and selecting Applications→Accessories→Terminal. Student Task 1: • Change to your home directory and install the training files with the script provided: sh > cd ˜ sh > /home/vlsi2/t1/install_t1 • Change to the design directory sh > cd training_1

The copied files and folders are arranged in a certain structure which is described in the next section.

3.1 Directory Structure The following figure shows the directory structure for a design directory that was created by the cockpit tool developed by the Design Zentrum (DZ) of ETH Zurich.

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design

.cockpitrc

Configuration for the cockpit

calibre

Final layout, DRC and LVS

docs

Links to documents

encounter

out

Final output files: netlist, layout, timing (Verilog,GDSII, SDF)

save

Save files for Encounter (Encounter native format)

scripts

Example scripts, run scripts (TCL)

src

Input source files: netlist, constraints, io placement

sample tech

Sample input files

Links to technology files, etc.

lef

Links to absracts and technology

lib

Links to timing libraries

modelsim

Simulation tool

simvectors

Stimuli and expected responses

sourcecode

VHDL sourcecode

synopsys

Synthesis environment

tetramax

Test vector generation, test coverage

In this structure, there are five subdirectories for C ADENCE S O C E NCOUNTER . It is strongly recommended to use them in the following way: out Place all final data to be exported from C ADENCE S O C E NCOUNTER in this directory. This includes the final netlist (the initial netlist gets modified by clock tree insertion, optimization etc.), layout and delay files that will be used for postlayout simulation and/or physical verification and chip finishing. A sample script that generates all these files is provided (scripts/exportall.tcl). save Put all C ADENCE S O C E NCOUNTER save files, i.e. files in native C ADENCE S O C E NCOUNTER format, in this directory. scripts Contains TCL scripts. By default several example scripts for common tasks are provided. It is highly recommended to develop a run script that contains all the commands used for your design. src All user input files should be placed here. These include the initial Verilog netlist, the I/O placement file, timing constraints file and clock tree definition file (all will be explained later in section 3.2). tech Holds links to technology specific files. Cockpit manages this directory automatically.

3.2 Input Files The input files required for back-end design with C ADENCE S O C E NCOUNTER can be divided into two categories: • Design files that describe (or are closely related with) the circuit, first of all the Verilog netlist of our synthesized design. • Technology files that describe the technology itself as well as libraries of standard building blocks implemented in this technology.

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Let’s start with the first category.

3.2.1 Verilog Netlist The Verilog netlist we obtain from synthesis contains standard cells, functional I/O pads and their interconnection information. While the functionality including scan circuitry is already complete, some special cells are still missing: • Supply pads to provide power and ground to the core (pads ’VCCKD’ and ’GNDKD’) and to the padframe (pads ’VCC3IOD’ and ’GNDIOD’). • Corner pads that need to be placed in the corners of the padframe to complete the power lines running inside the padframe (pad CORNERD). Due to the arrangement we have with our ASIC manufacturer, student designs are strictly limited in size. As a consequence at most 56 pads (not including the 4 corner pads) can be placed in the padframe. Furthermore, to ease chip testing on the ASIC tester two predefined power schemes have been established: 1. 40 signal pads, 16 supply pads Take a look at the following web page for an illustration of the power schemes and to obtain further information on constraints for the semester design projects. http://www.eda.ee.ethz.ch/index.php/UmcL180#Mini.40sic

With all this information we are now ready to add the missing corner and supply pads to our Verilog netlist. A typical Verilog netlist that you will obtain from S YNOPSYS D ESIGN C OMPILER will contain many levels of hierarchy. Each level of hierarchy is enclosed between the module name ( pin names separated by comma ) ... endmodule statements, where ’name’ refers to the name of the module (module is the Verilog equivalent of an entity in VHDL). In our case we need to add the pads to the top-level module which contains the rest of the I/O pads. The top-level design is almost always the last module definition in a Verilog file3 . Student Task 2: • Copy the Verilog netlist to encounter/src/ in order to have a clean copy of the initial netlist even if synthesis is rerun. sh > cd encounter/src/ sh > cp -p ../../synopsys/netlists/filter_chip.v \ filter_chip.v.initial The file specialpads.v contains four corner pads and 8 supply pads corresponding to the power scheme 1. As our design uses power scheme 1, no changes are required to this 3

The content of the module needs to be defined before it can be instantiated by a different module. Consequently the top-level module is the last to be defined, however not all Verilog files need to be hierarchical, a design can also be spread between multiple files

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file. For power scheme 2, we would have to comment out the eight additional supply pads (comments in Verilog start with //). What remains to do is to add the contents of specialpads.v at the right point, i.e. where the other pads are, to the initial netlist. • Using a text editor a , open filter chip.v.initial and find the definition of the top-level module ’chip’ by searching for: module chip Below this declaration you should see lines that instantiate the pads. Insert the contents of specialpads.v at this point. As long as you are in the module body, it does not matter where exactly you insert them. • Save the file as filter chip.v and exit the text editor. a

There are many text editors you can use. There are terminal based editors (vi, vim, nvi, joe, jed, pico, nano etc.), editors that are mainly terminal based but have a simple GUI (emacs, xemacs, gvim etc), and GUI based editors (mousepad, gedit, nedit, kate etc). Out of these emacs, vi (and derivatives), and nedit are the most advanced editors.

Remark: In the future you can use a small Perl script to add the specialpads to the initial netlist, i.e. sh > ./insert_specialpads ../../synopsys/netlists/filter_chip.v \ ./specialpads.v > filter_chip.v inserts the contents of specialpads.v into the last module defined in ../synopsys/netlists/filter chip.v and write the modified netlist to filter chip.v.

3.2.2 I/O File After the last step our Verilog netlist contains all pads. However there is no information that actually tells the tool where each pad should be placed. The pad placement is very important as it directly determines the PCB layout4 . In our case, we want all designs to share a common power and ground pad locations so that a single test board can be used on our ASIC tester. For practical reasons we have decided to use a 56-pin package for all designs. So even though the chip has only 48 physical pins, it will be placed in a package that contains 56 pins5 . Depending on the power configuration, a different bonding scheme will be used. These two configurations can be seen on the following webpage: http://www.eda.ee.ethz.ch/index.php/UmcL180#Mini.40sic

The cockpit will copy sample I/O files automatically to the src/sample directory6 . All lines starting with ‘#‘ are comments. The file consists of two main sections: globals and iopad. (globals [global definitions] 4 5 6

A good pinout could simplify the routing on the PCB, allow you to use fewer layers and result in less parasitics 8 pins will be left unconnected For this technology there will be four files. There will be two template files chip.io−template and chip−ep.io−template for the normal and extended power configuration respectively. These files have all the required power connections in place, and the data sections are commented out. There are also two example files that have fictional I/O placement where all pins are defined.

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) (iopad (topleft [pads that are on the top left] ) (left [pads that are on the left side] ) [definitions for other sides] ) For us the relevant part is the iopad section. This part contains eight subsections that define the names of the pad instances, and their locations in the four sides and four corners. We do not have to touch the corner specifications7 as they will be the same for all designs. We have to distribute the pads among the four sides of the chip top, right, bottom, left. If you look at the sample file you will see that for each pad there is a single line entry in the following form (inst name="NAME_OF_PAD"

offset=OFFSET_VALUE ) # pin no: PIN_NUMBER

The last part following # is a comment, it is there just for your information. Regardless of the power scheme you are using, we will use the same 56 pin package as illustrated in the webpage above. The PIN_NUMBER is just a reminder to show which particular location is being defined. The location is specified using the OFFSET_VALUE. C ADENCE S O C E NCOUNTER uses a coordinate system that bases the coordinate (0,0) on the bottomleft corner as shown in the figure below: top topleft

topright

1 2 3

left

right Side

Offset

0,0

bottomleft

bottom

bottomright

On the left and right side the pads will be ordered from bottom-to-top, and on the top and bottom side the pads will be ordered from left-to-right. This ordering can be quite confusing, as it is neither clockwise, nor counterclockwise. Therefore the aforementioned comments showing the actual pin numbers will be very useful. 7

topleft, topright, bottomleft, bottomright

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The OFFSET_VALUEs given in the template represent fixed locations for the given pad. It is very important that you do not change these values, as the chip-finishing part will rely on the pads being located exactly at these locations. You can assign your pads by writing the name of each pad into the corresponding NAME_OF_PAD. The name of the pad will be the name of the instance in the Verilog file. For example assume that you are using standard power scheme and your clock signal is assigned to a pad named pad_clock. In your Verilog file you would have the following entry for this pad: XMD ClkxCI_PAD ( .I(ClkxCI) [other pin definitions] ) If you now want to place this pad on pin number 54 of your package, you will find the subsection top in the I/O file and edit the line for pin 54: ... (iopad ... (top ... (inst name="ClkxCI_PAD" ... ) ... )

offset= 864.28 ) # pin no: 48

Be careful, do not modify the offset value while you are editing the I/O file. Since we use a fixed bonding scheme for the power and ground pins, all we need to do is extract the instance names for all our signal pads and place them by inserting within the appropriate inst name="" statement corresponding the OFFSET_VALUE which corresponds to the desired location. It is also recommended to put the clock pin (if possible) to pin number 48. All new test boards will make sure that the pin 48 has the best signal quality. Preparing the I/O file from scratch can be a lengthy and tedious task. To avoid unnecessary work during this exercise we will start with an almost complete I/O file, but before doing so we will describe the full procedure recommended when starting from scratch: 1. Start C ADENCE S O C E NCOUNTER and proceed to design import8 by selecting Design→\ Import Design. In this form make sure that the IO A SSIGNMENT F ILE is empty. 2. If everything works well, the design will be loaded. Now we can write out a template file that will contain all the names of the pads. Use Design→Save →I/O File ... to save an I/O file src/chip−sequence.io. You can select the SEQUENCE checkbox, however it is not imperative. What we need is only the names of the pads. 3. Copy the template I/O file src/sample/chip.io−template to src/chip.io. As noted earlier, this file includes all offset= statements, and all statements for corner and supply pads. 4. Using a text editor open the files src/chip.io and src/chip−sequence.io. You need to move the PAD_NAMEs from the file src/chip−sequence.io to the correct positions in the file src/chip.io. 5. All entries for data pins in the template file are by default commented out using ‘#‘ character. Do not forget to remove the comment character for the pads you are using. 8

Importing the design will be covered in detail in Chapter 4.

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Student Task 3: • Now, for this exercise you can start with the almost complete I/O file src/chip.io−incomplete\ instead of the template file. This file has all the pads placed properly with the exception of the 16 pads of the input bus DataInxDI which are still missing. Furthermore the file src/filter chip.sequence.io mentioned above has already been generated for you. The desired I/O assignment is depicted in the figure below and can also be found in the file src/filter chip.io.psa . • Create the complete I/O file and save it as src/filter chip.io. a

Postscript viewers were very common in the earlier days, you can use gv, kghostview, or evince to view this file

You can use the utility src/io2ps.pl to generate a postscript file from your I/O file. This utility will also verify if you have used the correct offset locations in you I/O file, and will report errors. For best results, you should also provide the Verilog netlist file, which will enable the script to make even more checks. sh > ./io2ps.pl filter_chip.io > filter_chip.pin_diagram.ps

DataInAckxSO_PAD

DataInReqxSI_PAD

pad_vcc_p4

ClkxCI_PAD

ScanEnxTI_PAD

pad_gnd_c4

RamTestxTI_PAD

pad_vcc_c4

DataOutxDO_PAD_14

DataOutxDO_PAD_15

DataOutxDO_PAD_13

ResetxRBI_PAD

pad_gnd_p4 1 2 3 4 5 6

pad_gnd_c3

34

DataOutxDO_PAD_6

33

DataOutxDO_PAD_5

32

DataOutxDO_PAD_4

31

DataOutxDO_PAD_3

30

DataOutxDO_PAD_2

29

pad_vcc_p3

15

16

17

18

19

20

21

22

23

24

25

26

27

28

DataInxDI_PAD_11

DataInxDI_PAD_13

DataInxDI_PAD_14

pad_gnd_c2

pad_vcc_c2

DataInxDI_PAD_15

DataOutAckxSI_PAD

DataOutReqxSO_PAD

DataOutxDO_PAD_0

DataOutxDO_PAD_1

pad_gnd_p2

pad_gnd_p1

DataInxDI_PAD_12

DataInxDI_PAD_0

DataInxDI_PAD_10

DataInxDI_PAD_1

pad_vcc_p2

DataInxDI_PAD_3 DataInxDI_PAD_2

14

7

43

8

44

9

45

10

46

11

47

12

48

13

49

35

DataInxDI_PAD_4

50

36

pad_vcc_c1

51

37

pad_gnd_c1

52

38

DataInxDI_PAD_5

53

39

DataInxDI_PAD_6

54

40

DataInxDI_PAD_8 DataInxDI_PAD_7

55

41

DataInxDI_PAD_9

56

42

pad_vcc_p1

DataOutxDO_PAD_12

The src/io2ps.pl utility uses a configuration file with the extension .pads. Per default the file src/io2ps\ .pads will be used. If you are planning to use the extended power scheme, you will have to add the configuration file src/io2ps−ep.pads to the command as well.

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pad_gnd_p3 DataOutxDO_PAD_11 DataOutxDO_PAD_10 DataOutxDO_PAD_9 DataOutxDO_PAD_8 DataOutxDO_PAD_7 pad_vcc_c3

3.2.3 Timing Constraints Just as for synthesis, we need to specify timing constraints for the backend design with C ADENCE S O C E NCOUNTER . With decreasing process geometries the impact of placement and routing on timing, power, etc. is steadily increasing. Therefore, timing analysis and optimization have become very important in order to arrive at a layout that (still) satisfies all requirements. As C ADENCE S O C E NCOUNTER supports most of the more common S YNOPSYS D ESIGN C OMPILER commands/constraints it should be rather straight forward to create an appropriate timing constraints file based on the constraints used for synthesis. Student Task 4: • There is an example constraint file src/sample/chip.sdc−sample that contains the most commonly used commands along with many useful and important comments. Copy this file to src/chip.sdc and modify it so that the following constraints get set (and nothing else!): – Define a 125 MHz clock – Specify 3.5 ns input delay for all inputs – Specify 5.0 ns output delay for all outputs – Specify an input transition time of 0.8 ns at all inputs – Specify a 15 pF output load for all outputs

3.2.4 Technology Files The tech directory and the two subdirectories contain technology files that describe the technology itself as well as libraries of standard building blocks implemented in this technology, i.e. standard cells, pads, RAM/ROM. • Technology files (UMCL180) lef/header6 V55.lef Base technology description, defines metal layers, vias, spacing rules, routing umcL180.capTbl Table used to extract parasitic capacitances and resistances for signal and power wires. streamout.map Layer mapping table used when exporting the final layout in GDSII format. • Library files (standard cells, pads, macro-cells) lef/*.lef Physical description, shape and allowed orientation of cells, layer and shape of pins, blockages, antenna information, ... lib/*.lib Functional description, timing and power information, maximum load/fanout or transitiontime allowed, ...

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3.2.5 Macro-cells The macro-cells for the umcL180 process are created using dedicated memory compilers. The specific memory compiler we have access to is able to create five different types of macro-cells with various capacities: • SU180 : single-port static RAM • SJ180 : dual-port9 static RAM • SY180 : single-port register-file10 • SZ180 : two-port11 register-file • SP180 : via programmable ROM The following parameters are used for the macro-cells: • words Number of words in the memory • sub-word size Number of bits within a sub-word of the memory. The sub-word is the smallest unit used for data access in the macro-cell12 . • number of sub-words per data word This parameter allows creating multiple sub-words. Each sub-word can be written to separately. For example, A 32-bit RAM can be configured as having a single 32-bit sub-word, or two 16-bit sub-words, four 8-bit sub-words and so on. • column or block multiplexer This parameter affects the geometry of the macro-block. This can have significant influence on the performance of the macro-block. There is no general rule to determine this parameter. Once the memory requirements are known, all possible geometries will be considered and the most suitable one will be determined. There are several available macro cells, their datasheets can be found under: /usr/pack/designkits-1.0-ma/umc_L180/faraday/gen/memaker/200901.1.1/datasheet.dz

If none of the available macro-cells suit your needs more can be easily generated on demand. Please contact the Microelectronics Design Center for this purpose. Our example design uses a single-port RAM named SY180_2048X16X1CM8. This RAM has 2048 words of 16-bits each (single sub-word) and a block multiplexer of 8. All necessary preparations to work with this macro-cell have already been done, so you do not need to do anything additional for this exercise.

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10 11

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dual-port memories have two completely independent access ports. At the same time two separate memory addresses can be accessed for both read and write. Although the name suggests that the memory is made out of individual registers, it is very similar in design to SRAM. In two-port memories, the read and write ports are separate, so you can simultaneously read and write. There are timing constraints for reads and writes to the same address, please refer to the memory compiler manual for details. In many places this sub-word is referred to as ’byte’. This might be slightly confusing, since a byte is commonly accepted to be an information unit consisting of 8-bits.

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4 Importing the Design Student Task 5: • Start C ADENCE S O C E NCOUNTER

a

either from your design directory by using cockpit

sh > cd ˜/training_1 sh > sh > icdesign umcL180 & • or from the encounter directory by issuing the command cd ˜/training_1/encounter cds_soc81 encounter a

This exercise uses version 8.1 of the Cadence SoC Encounter . There are newer versions of these software, however the main principles have not changed much so we will continue to use this version for this exercise, newer versions have slightly changed GUI elements, and improved capabilities for some functions.

We will now import our design. C ADENCE S O C E NCOUNTER uses a large configuration file that defines the design and technology files to be loaded as well as some global settings to be applied. Cockpit does automatically generate an appropriate sample configuration file src/sample/chip.conf that should be used to start with. Student Task 6: • Copy the sample file into the src directory. sh > cp src/sample/chip.conf src/filter_chip.conf • Select Design→Import Design ... to open the design import form. This form contains fields for all configuration options. At the bottom of this window, there are buttons to load and save the configuration from/to a file. Use the L OAD ... button to load the configuration file we have just copied to the src directory. • On the B ASIC tab make sure that V ERILOG N ETLIST :, T IMING C ONSTRAINT F ILE : and IO A SSIGNMENT F ILE : match your design. C OMMON T IMING L IBRARIES : and LEF F ILES : should already be correct. • On the A DVANCED tab the only setting you might want to adapt for your design is the D EFAULT D ELAY P IN L IMIT : in the category D ELAY C ALCULATION. We will explain this later. • Once you are happy with the configuration don’t forget to save your changes to the configuration file. • Click O K to import your design. Monitor the messages on the console for errorsa . Pay attention to the messages where the timing constraint files is loaded (“Reading timing constraint file”) to see if everything was accepted! If there are errors, you need to fix them! a

You can ignore warnings (SOCLF-58), (SOCLF-200), (TECHLIB-436), (SOCSYC-2), (EMS-27)

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We are now in the floorplan view of C ADENCE S O C E NCOUNTER which displays an empty floorplan with only the pads placed. All top level module(s) of the netlist are shown as a pink/purple square to the left and all macro-cells to the right. Note that all standard cells are inside the module(s).

5 Floorplanning Now we will have to decide how cells and macro-cells will be placed on our chip. This process is called floorplanning. For a standard design, our main concern would be to find a floorplan that will result in the smallest possible area, while fulfilling all performance and reliability requirements. This is purely driven by economical reasons, since chip costs are mainly determined by the area. In some cases there are additional geometrical constraints. The manufacturing company may impose certain

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limits to the aspect ratio of the final layout13 , or even dictate the maximum height or width of the layout. Back-end design is not only used for complete chips. Macro-cells that will be part of a larger systemon-chip design can also be designed in this way. In such cases there might be even more restrictions. For example, certain metal layers might be reserved for the system level. So the question is, “How small can my layout be so that I am still able to fulfill all specifications?”. As a lower bound, you will need enough area to place all your I/O pads and standard cells. Ideally, in terms of area (and assuming your design is not pad limited, see exercise 2), you will want to place standard cells without leaving extra space in between, completely filling out the core area. This is hardly ever possible because: • The number of interconnections that can pass through a certain area is limited by the number of metal layers available14 , wire width and minimum spacing requirements. Depending on the interconnection overhead, the area above the cells15 may not be sufficient for routing. • Timing is greatly affected by the placement of your cells. Placing them next to each other with no space in between not leave the tool any flexibility in placing cells. This in turn reduces the optimization options of the tool, like the ability to cluster cells that are closely interconnected. • All designs require power routing for operation. Some wires of the power connection limit where the cells can be placed, or restrict signal routing which in turn increases the area requirement. • The majority of designs require a clock tree to function. This clock tree is added during the backend design. This requires additional area for the buffers used in the clock tree. Furthermore, the clock tree synthesis algorithm can produce better results if it has more freedom to place its buffers. • Macro-cells, like the RAM in our example, usually require some extra space along the edges so that they can properly be connected to power and signal lines. • Designs that have a high switching activity require a lot of current for a short time which is called a surge. The power distribution network may need additional decoupling capacitors to store some charge that can provide some of the current of the standard cells during such a surge. Additional space for these decoupling cells may be required during placement. As a consequence, the standard cell rows (which form the core area) can not be filled completely with standard cells, in other words there needs to remain some free space in between cells. Utilization indicates to what amount the standard cell rows are filled. 100% utilization is the upper bound where all cells are abutted and there is no extra space, while a utilization of 50% means that half of the core area is empty. Usually, it is not possible to predict whether or not it is possible to fulfill all requirements with a certain utilization16 . You will have to try and find out. This is the main reason why back-end design is an iterative process17 . 13 14 15

16

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Especially in MPW runs, a lot of silicon area is wasted if all designs have wildly different dimensions. For our technology there are 6 metal layers. Cells in our technology use mostly the lowest metal layer Metal-1 and very rarely the Metal-2 for internal connections, all other layers are free for routing. Both placement and routing are separately NP complete problems, without completing the routing and placement you will not know if it is possible to fulfill the requirements. Obviously, technology plays an important role, and it is possible to give certain guidelines for a technology. However, backend design is always highly dependent on the design itself. You will usually see in a few iterations what is possible and what is not.

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5.1 Semester Projects The MPW provider used for the semester projects offers modules caled Mini Asic (mini@sic) with a size of 1519.62 µm × 1519.62 µm. Therefore, the chip size for the semester project ASICs is fixed. Please refer to the following web page to learn the details. http://www.eda.ee.ethz.ch/index.php/UmcL180#Mini.40sic

As a consequence, we only have to make sure that our design fits on this area, and there is no need to find the smallest possible layout. We may however need to constrain the core area to make it smaller if the utilization is too low, since a spread out design has longer interconnections that may adversely affect timing.

5.2 Sketching a Floorplan Before we go on with C ADENCE S O C E NCOUNTER we need to make some planning and understand some key concepts. The figure on the following page is an example floorplan (not an ideal one) that shows the important concepts. In C ADENCE S O C E NCOUNTER die area corresponds to the total silicon area available to place pads (excluding bonding area for this technology) and core cells. For the semester projects this is strictly limited to 1519.62 µm × 1519.62 µm. All pads (I/O, power and corner) are placed in what is known as the padframe. The remaining area can be used for the core of the chip. For semester projects the theoretical maximum for core area is 1239.38 µm × 1239.38 µm = 1.54 mm2 . As can be seen from the figure, the core area is surrounded by a core power ring. In its simplest form this consists of two (one for VCC, one for GND) wide18 metal lines that evenly distribute the power all around the chip. In order to leave room for the power ring, we need to leave a certain I/O to core spacing. The standard cells are designed in such a way that, when placed next to each other their VCC and GND pins can be connected with a horizontal power line. These horizontal lines are then extended to the core power ring. These power connections are relatively narrow (0.76 µm in the technology that we use) and run over the entire width of the core area. This could be a problem for designs that consume much power, since the cells towards the middle would not have a good power connection19 . To improve this, vertical power stripes that connect to the horizontal power lines can be added, thereby forming sort of a mesh. The core area is filled with standard cell rows on which later all standard cells will be placed. In the same area we will usually also need to make room for our macro-cells. Most macro-cells need some free space around themselves. This free space is required to make signal connections, add a block power ring around the macro-cell or simply to prevent standard cells from being placed too close to the macro-cell. We will define a block halo to specify this free space.

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The width of the metal line depends on the amount of current drawn from the line, you will be able to judge this better after exercise 3 which is dedicated to estimating the power consumption. We will mostly use a width of 20 µm, since this is the widest metal that can be manufactured without slotting (wider metal lines require slots/holes which break up the metal shape). The problem is that if much current is drawn, there will be a significant IR drop along the power lines. The cells in the middle will be supplied with a lower VCC than the ones on the sides. This could dramatically effect the performance of the system.

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When placing a macro-cell, you should also take into account where the power and signal pins of the block are located and what metal layer they are on. Often signal connections are only on two edges and you want them to face the core and not the I/O pads. Now, when we consider all the above, the core area that remains free to place core cells on is much smaller than the 1.54 mm2 that we started with. Our example design has a total cell area (including RAM) of 0.82 mm2 and should therefore comfortably fit into the designated area. 1519.62 µm

VDD GND Power Stripe Standard Cell Power Connections

Block Power Ring Standard Cell Row

1239.38 µm

Macro Cell (RAM) Standard Cells

I/O and Corner Pads Placed on the Padframe

Core Power Ring

Block Power Connection

Power Pad Connections

Block Halo

I/O to Core Spacing

5.3 Initialize Floorplan We are now ready to proceed with C ADENCE S O C E NCOUNTER . Student Task 7: • From the menu select Floorplan→Specify Floorplan.... A large window will open. • Select the D IE S IZE BY: W IDTH AND H EIGHT option and make sure that both values are 1519.62. • Now we need to specify the I/O to core spacing by filling in the four values under the C ORE M ARGINS BY: entry. There must be sufficient room for the power ring around the core area.

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Larger values will reduce the area available to place the core cells thereby increasing core utilization. As noted earlier, some iterations are usually required to find optimal values for a particular design. • In this exercise we will assume that we will use one VCC and one GND line of maximum width 20 µm. We need some extra space between the lines and, for the moment, we can start with a distance of 50 µm for all sides and click on OK.

The floorplan should now look like shown in the screen-shot below. Note that the pads are all placed at their proper locations as the I/O file used during design import specifies absolute locations and we made sure that the die size stays fixed to the proper size during the initialize floorplan step. Student Task 8: • Next we need to place the RAM macro-cell. Change the cursor mode to M OVE /R ESIZE /R E SHAPE by selecting the appropriate icon (next to the ruler icon) or use the keyboard shortcut ’SHIFT-R’. Now you can select the RAM macro-cell and drag it to any location you like. The blue lines displayed are so called flightlines that show where the signal connections to the block are. You can change the orientation of the RAM by either using Floorplan →Edit Floorplan→\ Flip/Rotate Instances ... (or press ’r’), or with the attribute editor (press ’q’). Note that the RAM macro will completely block Metal-1, Metal-2, Metal-3 and Metal-4. Only Metal-5, Metal-6 will be available for routing over the RAM macro-cell20 .

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By default, the internal structures within a cell or block are not displayed. You need to make “Cell Blkg” visible to see the so called blockages within a cell.

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5.4 Power Planning The next step is to create the power distribution network. The Verilog netlist that we started with does not contain any power connections, therefore we need to create this connectivity now. We have to connect the power/ground pins of all instances to the respective global power/ground net that was specified on the D ESIGN I MPORT form (category P OWER on the A DVANCEDtab)21 . This can be done using the Floorplan →Connect Global Nets ... form or you can use the globalnet.tcl script provided. Student Task 9: • Execute the script provided by typing on the command line of C ADENCE S O C E NCOUNTER (not GUI): enc > source scripts/globalnet.tcl

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There is also a special rule required if there are logic one/zero values 1’b1/1’b0 instead of TIE1/TIE0 cells in your netlist. You should however not have such logic values in your netlist.

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Next we will add the core power rings that distribute power all around the core. Student Task 10: • Select the menu Power →Power Planning →Add Rings.... A large window will appear. The N ET ( S ) field on the top defines for which nets rings will be created. The default is to create power VCC as well as ground GND rings. • In the R ING C ONFIGURATION section you can specify on what layers the ring segments will be created. Select metal5 H for TOP and B OTTOM and metal6 V for L EFT and R IGHT. Specify W IDTH as 20 µm, S PACING as 1.5 µm and O FFSET as 4 µm and click O K.

There are many alternative power distribution schemes that can be used. The one that we have chosen here is a very simple one. We have selected the upper metal layers Metal-5 and Metal-6 for the ring, because in this technology Metal-6 is thicker and consequently has less parasitic resistance which is desirable for power distribution. For your own designs, you should perform a power analysis (topic of Training 2) to find out the best power distribution approach that matches your design. The width has been chosen as 20 µm for convenience reasons. Basically the wider the power connection, the better. But as already mentioned earlier, in this technology, metal lines wider than 20 µm need to be slotted (’stress relief slots’) which requires extra effort. As an alternative to slotting it is also possible to create several smaller parallel rings, e.g. two VCC and two GND rings.

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S PACING determines the distance between the two nets and O FFSET determines the distance between the core area and the innermost ring. We also need a (partial) ring around the macro-cell, you will see later why this is necessary. Student Task 11: • Select the menu Power→Power Planning →Add Rings... just like before. This time in the R ING T YPE box, select B LOCK RING ( S ) AROUND. You can leave the selection at E ACH BLOCK since we have only one block anyway. C ADENCE S O C E NCOUNTER is usually smart enough to create wires only on the edges where no power lines are yet, i.e. to not create new wires on top of the core ring. • If this fails you can specify the segments and connections you want on the A DVANCED tab. • Fill in the values/settings similar to that of the A DD R INGS and click on O K. At any point if you wish to delete part of the floorplan you can: • use the U NDO feature by simply pressing ’u’ • select and remove objects of a specific class (press ’d’) • use the menu option Floorplan →Edit Floorplan →Clear Floorplan... • select an object and hit the ‘Del’ key on the keyboard Student Task 12: • Also, you can save or load (restore) your floorplan at any time using the menu Design →Save →Floorplan ... and Design →Load →Floorplan ... respectively. • Save your floorplan to the save directory. At this point power is to the standard cells arrives from the sides. Especially for fast designs the standard cells in the middle of the standard cell row will not receive sufficient power it is important to add vertical stripes to improve the power distribution. Student Task 13: • Select Power →Power Planning →Add Stripes .... The S ET C ONFIGURATION part of the window defines the properties of one stripe set. The S ET PATTERN part defines how many stripes will be added. We can either choose to insert a fixed number of sets or only specify the distance between two sets S ET- TO - SET DISTANCE : • In the F IRST /L AST S TRIPE part, we select R ELATIVE FROM CORE OR SELECTED AREA. Add to X FROM LEFT and X FROM RIGHT a value stripe sets in such a way that the standard cell rows get divided into three equally long pieces. See the screen shot for width, spacing and layer. Note: You can fine tune this later by moving the stripe sets. • By default stripes will continue over macro cells. To prevent this, select the O MIT STRIPES INSIDE BLOCK RINGS option in the S TRIPE B REAKING section of the A DVANCED tab.

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It is rather easy to move wires in C ADENCE S O C E NCOUNTER . Click on the move wires button (or press ’m’), select the wires you want to move, and drag them to their new location. C ADENCE S O C E NCOUNTER will make sure that electrical connections remain intact. If you want you can use this to fine tune the stripe placement. We still need to define a block halo for the RAM macro-cell. This is necessary to keep standard cells from being placed to close to the RAM and also to avoid problems when routing the power lines of the standard cell rows. The figure below illustrates one common problem with the block halo.

Terminated Power Line (good) Standard Cell Row Macro-Block

Dangling Power Line (bad)

Power Rails

Standard Cell Row

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Block Halo

In this figure, only two standard cell rows are shown. The block halo around the first row extends far enough to cover the two power lines22 . This is like it should be. For the second row, the block halo does not cover the power rails, and when making the power connections C ADENCE S O C E NCOUNTER will try to extend the power connection past the power rails as shown in the figure. This leaves a dangling power line23 . While this will not render your chip useless, it should be avoided. Student Task 14: • From the menu select Floorplan →Edit Floorplan →Edit Halo.... A window will appear, where you can specify a keep-out zone for routing and/or placement around the macro-cell. Usually we only need a Placement Halo. The size will depend on your power routing/floorplan. • Create an appropriate Placement Halo.

Notice that the I/O pads are placed with some distance between them24 . At some point in the design flow we need to close the gaps between the I/O pads in order to complete the supply rings that run around the core (within the pad cells) and are required to supply the circuitry within of the pad cells. Student Task 15: • Instead of using wires, we will place so called filler cells that completely fill the gaps and establish the required connectivity. There is a script that will automatically insert matching filler cells. Type the following in the C ADENCE S O C E NCOUNTER console window enc > source scripts/fillperi.tcl

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This is just for illustration. It is not possible to draw a block halo that has this (L) shape. This sort of dangling wires are known as geometry antenna in Cadence SoC Encounter This is due to the contraints set by the company that bonds the chips. They specify that the minimum distance between two adjacent pads can be 90 µm. Since even a core-limited pad in this technology is roughly 60 µ wide, we need to place them with gaps in between.

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Now we need to finalize the power connections of the chip. The following connections still need to be made: • The core ring needs to be connected to the core supply pads (VCC3IOD and GNDIOD). • All standard cells need to be connected to VCC and GND lines. • All macro-cells need to be connected to VCC and GND lines. Student Task 16: • Select Route →Special Route ... from the menu. SRoute is the special net router, and is only used to make power connections. The R OUTE : part contains the different connection types we have listed above. B LOCK PINS are macro-cell power connections, PAD PINS are the connections from the core supply pads to the core ring. We will not need PAD RINGS since we have already used filler cells to complete these rings. S TANDARD CELL PINS will add power lines to the standard cell rows. Finally, if you still have stripes that are not connected to power (not very likely) you can use the S TRIPES ( UNCONNECTED ) option. • While it is possible to route all connections at the same time, it is strongly recommended to do it one by one: 1. Start with PAD PINS. If nothing happens you have most likely forgotten to source the globalnet.tcl script. 2. Route B LOCK PINS. Check the result, did the router connect the macro-cell the way you wanted? If not you may need to study the A DVANCED tab of the SRoute window. If all fails you can edit the connections manually. 3. Route the S TANDARD CELL PINS. This should create many horizontal Metal-1 lines that connect to the rings and stripes. Look for dangling wires around the block halo (adjust the block halo if necessary). We are now finished with floorplanning. Your floorplan should look similar to the following screen shot.

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6 Placement We will now start with the placement of the standard cells in the core area. Placement is a very computation intensive problem, and mostly heuristic algorithms are used for this purpose. Student Task 17: • Select Place →Standard Cells.. .... We want run a full placement and not an incremental or just the quick prototyping one. I NCLUDE P RE -P LACE O PTIMIZATION however is very useful as it removes all buffers/inverters trees from the netlist which will help us for timing analysis as you will see later. • To set advanced options click M ODE. Set C ONGESTION E FFORT to L OW and deselect RUN T IMING D RIVEN P LACEMENT as timing driven takes much longer and might not help that much to improve timing. There are several other options that you can set, but at this time we will leave them as they are. Apply the changes by pressing OK • You will come back to the placement window seen below, click OK to start placement. This may take some time. We have to warn you about the various performance related options such as C ONGESTION E FFORT and RUN T IMING D RIVEN P LACEMENT above. In the exercises sometimes we will advise you to use certain settings for these options in order to reduce runtime, or because for this particular design we have found out that a particular option gives better results. When you do your own designs, you

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should consider evaluating which options are better suited rather than copying all options from this exercise.

For each standard cell, the placement algorithm will try to find the optimum location so that there is a feasible routing solution and the total length of the connections is minimized. Examine the placement by using the design browser (switch to the physical view). You will notice that standard cells within the same entity are mostly placed next to each other. The available space and the placement of macro-cells and I/O pads can have a great influence on the placement of standard cells. Even though more space seems to be a good idea, too much space sometimes results in placements where the average distance between standard cells and consequently the delays caused by wire capacitance/resistance become larger. Only experience and several iterations will allow you to find a placement for your circuit that is close to optimal. Note: Visibility of S PECIAL N ET is turned off in the next screen shot.

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The results for placement (and later routing) are strongly design dependent. For example, structures with many interconnections such as look-up tables will usually need much more space than synthesis predicted as the cells need to be spread out in order to have enough space to route all the interconnections. This is why generalizations for back-end design, such as ”During back-end design, your circuit area will increase by 10%” don’t work very well. Student Task 18: • Let us save the entire design with Design →Save Design As →SoCE. This will save the configuration file, netlist, floorplan, special route, placement and routing files as well as the current mode, options and preferences. A design saved in this way can be restores using Design →Restore Design ... →SoCE. The space required is surprisingly small as most files are compressed and the library files do not get saved along with the design. • Remember to save under the save directory. Alternatively you could also just save the placement. Select Design →Save →Place \ .... During synthesis, S YNOPSYS D ESIGN C OMPILER assigns constant logic values to two special standard cells named TIE0x and TIE1x, where x is a drive strength modifier. This creates a small inconvenience, as often one of these cells is assigned to drive many outputs at the same time, creating relatively long interconnections. There is sufficient place on the chip to place several of these cells. We will use a script that first removes all these cells. Then we will set the rules for placing these cells. The example script scripts\ /tiehilo.tcl sets the maximum number of connections driven by a single cell to 20, and the maximum distance between the pin and the tie cell to 250 µm. And finally we insert the tie cells according to the rules we have defined. Student Task 19: • At the command line type: enc > source scripts/tiehilo.tcl

7 Timing The synthesis tools we currently use for HDL synthesis (S YNOPSYS D ESIGN C OMPILER) are not aware of any instance placement information. Therefore the interconnects can only be estimated based on a statistical model, i.e. the fanout of a net determines its length, capacitance, resistance and area. Now that the placement and even trial-routing is available the timing might differ considerably from the numbers obtained from S YNOPSYS D ESIGN C OMPILER.

7.1 Analysis C ADENCE S O C E NCOUNTER has a practical timing analysis function, where you only have to specify the state of the design (see below) and the A NALYSIS T YPE (Setup or Hold) you want to run.

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Pre-Place design is not placed Pre-CTS design is placed but clock tree is not yet inserted Post-CTS design is placed and the clock tree is inserted Post-Route design is placed and routed Sign-Off will use extra tools for even more precise analysis. We will not use this as these tools are not installed/setup. Depending on this state, trial route (a very simple, but fast routing) and/or parasitic extraction might be run automatically prior to the timing analysis. This will improve the accuracy and help to avoid unnecessary iterations. Student Task 20: • Open Timing →Analyze Timing and make sure P RE -CTS and S ETUP is selected. • Start the timing analysis by clicking O K. Note: You could also do this from the command line with enc > timeDesign -preCTS As the design is not routed, C ADENCE S O C E NCOUNTER will perform trial route and parasitic extraction before doing the timing analysis. A short summary will be displayed on the console (the actual numbers may differ slightly): +--------------------+---------+---------+---------+---------+---------+---------+ | Setup mode | all | reg2reg | in2reg | reg2out | in2out | clkgate | +--------------------+---------+---------+---------+---------+---------+---------+ | WNS (ns):| -9.069 | -6.554 | -9.069 | -0.686 | -7.328 | N/A | | TNS (ns):| -2684.3 | -1776.9 | -2392.1 | -1.172 | -43.761 | N/A | | Violating Paths:| 861 | 732 | 454 | 7 | 6 | N/A | | All Paths:| 1807 | 1342 | 817 | 18 | 6 | N/A | +--------------------+---------+---------+---------+---------+---------+---------+ +----------------+-------------------------------+------------------+ | | Real | Total | | DRVs +------------------+------------+------------------| | | Nr nets(terms) | Worst Vio | Nr nets(terms) | +----------------+------------------+------------+------------------+ | max_cap | 187 (187) | -3.774 | 188 (188) | | max_tran | 368 (13826) | -8.333 | 387 (13867) | | max_fanout | 0 (0) | 0 | 0 (0) | +----------------+------------------+------------+------------------+ Density: 59.566% Routing Overflow: 0.00% H and 0.25% V ------------------------------------------------------------

The summary gives a very good overview of the current design timing. Some explanations: • The analysis was run in setup mode, i.e. setup time checks were performed but no hold time checks.

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• The columns contain numbers for all path in the design (ALL) or for specific path groups, e.g. reg2reg for all register to register paths. • Worst negative slack (WNS) reports the slack for the most critical path. Negative numbers mean that the constraints are violated by this value. • Total negative slack (TNS) is the sum of WNS for all violating paths. Together with the number of violating paths this figure helps to see how severe the violations are. • Real/Total DRV show (electrical) design rule violations, some libraries have a maximum transition time for all nets. The report above shows that 370 nets have a transition violation (the signal takes too long to change from logic-1 to logic-0 or vice versa). In addition 135 nets have a maximum capacitance violation (the total amount of capacitance driven by a net exceeds the limit set by the design library). These violations are mostly related to excessive parasitic capacitance due to interconnections, and generally cause timing violations as well. However, even if a DRV does not cause a timing violation it needs to be fixed. • D ENSITY and R OUTING OVERFLOW show the placement utilization and routing resources, i.e. are a measure for the feasibility of the current floorplan/placement. Remark: Refer to exercise 4 of VLSI I25 if you have problems with timing concepts. The summary looks really terrible. Obviously we have many timing violations that we need to have a closer look at, before we try to optimize the timing with C ADENCE S O C E NCOUNTER . Here are some important points to consider when doing so: • The timing depends entirely on the constraints you have specified in the file src/chip.sdc. The most common mistake is to have errors in this file. Before you go any further make sure that your timing constraints are correct. • Make sure to not accidentally use constraints that were written for the core level (chip without pads) at the chip level (with pads) and vice versa. The pads affect the I/O timing quite a bit and the drive capabilities of a standard cell and an output pad are entirely different, i.e. set_load needs to be very different. • Inputs and outputs used for test and debugging may cause timing violations. Most of these signals are not dynamic (they are not toggled during normal operation) and the timing paths originating from these inputs or ending at these outputs should be ignored, i.e. left unconstrained or explicitly disabled. • To speed up delay calculation C ADENCE S O C E NCOUNTER does not compute the timing of nets with a fanout above a certain limit but rather swaps in predefined values for delay, capacitance and transition time. All these numbers are specified on the D ESIGN I MPORT form on the A DVANCED tab in the ”Delay Calculation” category. As a result you will not see the real timing26 of these net in timing analysis and furthermore optimization will not see (and therefore not fix) violations27 on these nets. However, this is usually the desired behavior as we give these nets a special treatment anyway (with CTS).

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You can access the exercise descriptions, files, and solutions under /home/vlsi1/u4. To see the real timing you can change the limit on-the-fly from 1000 to a very high value in the console with setUseDefaultDelayLimit 100000. More on this topic later. DRV violations will be fixed but no setup/hold violations. Clock nets are even more special, also no DRV fixing will be done there.

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Let’s now examine the detailed reports that were generated by timing analysis and can be found in the timingReports folder. Each analysis produces multiple files. Among these there are three files dedicated to design rule violations (max capacitance: *.cap , max fanout: *.fanout, max transition time: *.tran violations), and separate *.tarpt timing analysis report files for different path groups (in2out, in2reg, reg2reg, reg2out) Student Task 21: • Where do the violating paths in the in2out path category start? • Where do the violating paths in the in2reg path category start? • Do the paths in reg2out and reg2reg look like normal path that should be optimized to meet timing or is there something wrong? • Why are the reg2reg paths too slow? Look for large numbers in the Delay column and check the drive strength of the corresponding cell. There are several different problems in the .sdc file that we have used. First of all, two of our inputs should not be considered for timing analysis28 . We also have several nets (clock, reset and scan enable) that we will take care of separately (using the clock tree synthesizer, which we will see later). These nets will show up in the DRV reports. We do not want to solve timing related problems for these nets (since they will anyway be solved later), the time and effort required to optimize these nets could prevent other parts of the design to be optimized. We can use the D EFAULT P IN L IMIT feature of C ADENCE S O C E NCOUNTER to stop C ADENCE S O C E NCOUNTER from extracting timing information (and reporting timing violations) for the nets that we will be optimizing later on. By default the pin limit of C ADENCE S O C E NCOUNTER is set to 1000. In our case this number is too high (we have slightly more than 400 flip flops in our design). Student Task 22: • Let us see the nets which have a large fanout. Report all nets with e.g. more than 400 pins. Use the console command: enc > report_net -min_fanout 400 • Now set a suitable limit with the command enc > setUseDefaultDelayLimit so that the high fanout nets will not be considered for timing. Also make the necessary changes to the timing constraints file src/chip.sdc to disable the offending inputports. Reload the timing constraints by selecting the menu Timing →Load Timing \ Constraint .... • Then rerun timing analysis. If you have done everything correct, the only setup violations should be in the path group registerto-register and register-to-out. There should no longer be pins that belong to scan enable or reset network in the transition time violation report. 28

Cadence SoC Encounter provides a special timing calculation mode that is called Multi-Mode Multi-Corner Analysis (MMMC). In this mode it is possible to define several scenarios (i.e. separate test and functional modes). The setup for MMMC is slightly involved and will not be covered as part of this exercise.

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7.2 Optimization In order to (better) meet the constraints, C ADENCE S O C E NCOUNTER can try to optimize the design at every stage of the design process. In our case, the worst setup time violation is about 5.8 ns (for a 8 ns period), although the netlist delivered by the synthesis tool had no timing violations. This is due to differences in interconnect parasitics between the two tools. While the synthesis tool relies on an estimate (statistical model based) C ADENCE S O C E NCOUNTER can use the real placement and (trial-)routing at hand. Consider the following line from a timing report (broken down over many lines for readability) Path 1: VIOLATED Setup Check with Pin i_filter_top/u_filter/u_filter_stage_5/ RegxDP_reg_42_/CK Endpoint: i_filter_top/u_filter/u_filter_stage_5/RegxDP_reg_42_/D (ˆ) checked with leading edge of ’ClkxCI’ Beginpoint: i_filter_top/u_ram_wrapper/i_ram/DO5 (ˆ) triggered by leading edge of ’ClkxCI’ Path Groups: {reg2reg} Other End Arrival Time 0.000 - Setup 0.149 + Phase Shift 8.000 = Required Time 7.851 - Arrival Time 14.405 = Slack Time -6.554 Clock Rise Edge 0.000 = Beginpoint Arrival Time 0.000 Timing Path: +----------------------------------------------------------------------------------------------------------+ | Instance | Arc | Cell | Slew | Load | Delay | Arrival | | | | | | | | Time | |-----------------------------------+---------------+--------------------+-------+-------+-------+---------| | | ClkxCI ˆ | | 0.000 | 1.828 | | 0.000 | | ClkxCI_PAD | I ˆ -> O ˆ | XMD | 0.000 | 0.000 | 0.000 | 0.000 | | i_filter_top/u_ram_wrapper/i_ram | CK ˆ -> DO5 ˆ | SY180_2048X16X1CM8 | 0.130 | 0.033 | 1.750 | 1.750 | | i_filter_top/u_ram_wrapper/i_test_| A ˆ -> O ˆ | MUX2 | 8.441 | 1.874 | 3.973 | 5.722 | | bypass_mux5 | | | | | | |

The last line reports an standard cell instance MUX2 with low driving capability (2) that has to drive a big load on its output (1.876 pF). The propagation delay is therefore huge (3.98 ns). The timing of the same cell as reported by synthesis are: Delay: 0.15 ns, Slew: 0.09, Load: 0.01. While this is an extreme case you see how synthesis can be wrong without knowing the actual placement and wire loads. Student Task 23: • Open the optimization form by selecting Timing →Optimize .... D ESIGN S TAGE needs to be set to the current design stage. Some options are only available for certain stages, e.g. hold time optimization can not be performed during PRE -CTS as it doesn’t make much sense. Timing is not the only thing that can optimized. Most technologies specify design rules like maximum transition time, maximum capacitance driven by a certain cell or maximum fanout. • After pressing the M ODE button, within the T HRESHOLDS section you can find options that can be used to tighten the constraints in order to get some margina .

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• Set the options as shown in the figure below and hit OK. Watch the progress of the optimization in the console window. C ADENCE S O C E NCOUNTER is very verbose with its actions. a

Cadence SoC Encounter will already automatically add a small margin on its own (internally)

During optimization C ADENCE S O C E NCOUNTER can select different drive strengths for cells, add/remove buffers and inverters, move instances or even restructure part of the logic (just like synthesis does). Optimization is done using iterations of timing analysis, optimization, trial-route and parasitic extraction. As a last step C ADENCE S O C E NCOUNTER performs a timing analysis on the optimized design, prints the summary to the console and writes the detailed reports to the timingReports directory. Student Task 24: • Take a look at the summary and the final reports generated. There should be no violations left. But what happens if we can not fix the violations with optimization? Again, first make sure to understand what your constraints are and why they are violated. Often there are errors in converting the design specifications to constraints (is the input delay really 3.5 ns? Also for this pin?) and describing them properly with the commands available. If you still have problems, there are three levels where you can reach a solution: • Optimization during backend design (C ADENCE S O C E NCOUNTER ) C ADENCE S O C E NCOUNTER can optimize the design at every stage of the design process. In general, the earlier the stage, the more changes can be done, e.g. P RE -CTS optimization has much more flexibility than P OST-R OUTE optimization. At the P RE -CTS stage registers can be moved and resized, this will no longer be possible after clock tree insertion. On the other hand, the parasitic interconnect information is much more accurate with later stages of design, so the timing information (and hence the optimization goals) will be more accurate. We can (re)run the optimization at various stages, try a new placement or even start with a new floorplan. It is impossible to give general guidelines, you will have to see what works best for your design. If you are far from meeting your target (e.g. for a 10 ns clock, if after all optimizations you still have a timing violation of 2 ns), you may need to go back to synthesis.

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• Optimization during synthesis Once you have tried to place and route a netlist you will get a better idea about the relationship between synthesis results and back-end results (area and timing wise). You may use this information to adjust the timing constraints and re-synthesize the circuit. • Architectural optimizations If nothing else helps, you will have to modify your architecture. During this iteration you will have a much better idea about what is critical for your circuit. If all of the above fails, you will have to see if the specifications could be changed. Student Task 25: • Your design has changed considerably as the optimization algorithms have modified the netlist and placement. Save it by using Design →Save Design As.

8 Clock Tree Insertion The fan-out of a net refers to the number of inputs driven by a particular output. High fan-out nets (that drive hundreds or even thousands of inputs) need to be handled differently from standard interconnections. Note: For timing analysis we did adjust the pin limit (setUseDefaultDelayLimit) in order to treat them differently. Every synchronous circuit has at least one high fan-out net, namely the clock net. For most circuits reset and scan-enable signals have to be distributed to each and every flip-flop as well. The main problem with high fan-out nets is the large load capacitance that needs to be driven. Each driven input adds its own input capacitance to the total load capacitance and in addition, the interconnection required to distribute the signal to all these inputs increases the load capacitance further. There are three important parameters for such nets: Transition time This is the time it takes to change the logic level of a node (e.g. 0 → 1). Basically, the more load an output has to drive, the more time is required to charge this load. CMOS drivers consume additional short circuit current during the transition, therefore long transition times are not very welcome. Furthermore, noise on signals with long transition times can result in glitching. Most libraries set an upper limit for the transition time (for the technology we are using this is 1.79 ns for typical libraries). To lower the transition time, a tree of buffers can be inserted so that the total load is shared between the buffers. The lower the desired transition time, the more buffers are required. Insertion delay The time required for the signal to travel from the driver to the end-points. This delay is usually different for each end-point. Each level of buffers in the buffer tree will add a delay to the signal. Skew The difference between insertion delays of different end-points. To minimize skew, a balanced buffer tree has to be built. Generally, the lower the desired skew the more buffers are required. What parameters are most important depends on the type of net:

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Clock Our main concern is to reduce the skew, since it will effect our timing. The maximum skew depends on the clock period. As an example, for a 20 MHz clock a clock skew of 0.5 ns is acceptable. But for a 200 MHz clock, the same skew equals to 10% of the clock period and would be to high. If you over-constrain your skew, you will need a deep (and large) clock tree and your insertion time will rise, which will affect your input and output timing. Therefore you will want to balance the skew against insertion delay and the number of buffers. Constraining maximum insertion delay too low will usually degrade results. Usually, a tree that gives you an acceptable skew will also give you a decent transition time, so you don’t have to worry about that. Reset We are interested in propagating the reset within one clock cycle to all flip-flops in our design. For designs with on-chip reset synchronization this is strictly required. The insertion delay should therefore be less than the clock period, transition times within the bounds imposed by the technology and skew doesn’t matter at all. Scan Enable Very similar to the reset signal. Usually a slower clock is used for scan testing, therefore we can allow even a larger insertion delay. For transition time and skew the same holds true as for the reset. Sink Tran Buf Tran Sink Tran Buf Tran AutoCTS Root Pin Sink Tran Buf Tran Sink Tran

Min Delay Max Delay Max Skew

In C ADENCE S O C E NCOUNTER , clock tree synthesis (CTS) is used to generate optimized buffer trees to drive high fan-out nets. It can be configured to satisfy a variety of constraints. Student Task 26: • A sample clock tree synthesis configuration file can be found under src/sample/chip.ctstch\ −sample. The sample file contains three different configurations for a clock, a reset and a scan enable signal. • Copy this file to the src directory and adapt the ’AutoCTSRootPin’ statements to match your design. • For educational purposes, change the clock tree specifications as follows: max. skew 0.2 ns, max. insertion delay 4 ns, max. transition time at buffers 0.6 ns and at clock pins 0.4 nsa

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Take a closer look at the other two trees too. a

It is usually not a good idea to specify a small max. insertion time such that this becomes a limiting factor for CTS. Results may degrade significantly and for most designs the insertion delay is not very important anyway.

If the design employs a reset synchronization register (the example design has one) the source of the reset tree must be the output of the synchronization register. Note that there is a special option named SetASyncSRPinAsSync YES for the reset tree definition. This allows set and reset pins to be considered as targets for the clock tree optimization. The scan-enable signal is also a special case. Normally the clock tree synthesis algorithm starts at the AutoCTSRootPin and traces through the netlist in order to find valid endpoints. Per default, combinational gates will be traced through and clock and asynchronous input pins of sequential elements (flip-flops) will be stopped at. By specifying the NoGating rising option, we can make the tracer stop at the first gate encountered. This is necessary since the scan enable signal is often connected to multiplexers and we want their input pins to be endpoints. Once this option is underway you need to specify the internal pin of the pad driving the scan-enable signal, otherwise tracing will stop prematurely at the pad cell. Student Task 27: • Read in the clock tree specification by selecting Clock →Design Clock ... from the menu. Using the browser select the clock tree specification file you have just modified. Press L OAD S PEC. DON’T PRESS OK yeta . You should now see a summary for all three clock specifications on the console, check it. • Our netlist may have some buffers on the high fan-out nets we want to build trees on. We need to remove them prior to CTS with the following command: enc > deleteClockTree -all a

Pressing OK will start the clock tree insertion. We need to make sure that the clock tree specification is correct before we go ahead with this step. If you accidentally pressed OK here, it is advised to restart from the last saved point.

A large number of errors can be discovered by analyzing the pins connected to these nets, even before building a clock tree. Student Task 28: • Select Clock →Trace Pre-CTS Clock Tree .... To start the trace, click on the icon on the top left and accept the default trace file name. A summary will be displayed on the console and the content of the trace file visualized in the GUI.

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We can see how the trees currently look like and what pins are connected to them. Look also at the trace file directly. Things to look for include: • Clock, reset, or scan-enable connecting to unexpected input pins, e.g. the reset signal should not connect to pins other than asynchronous set/reset pins of sequential elements. • Unexpected latches on the clock tree can be discovered this way (G or GB pin). • Discrepancy between the number of endpoints of clock, reset and scan trees. For our example numbers are as follows: – clock tree: 443 with 442 flip-flop CK pins + 1 RAM CK pin – reset tree: 441 flip-flop RB pins – scan tree: 447 with 441 flip-flop SEL pins + 6 mux S pins, to choose between the functional and test (scan chain) output signal. As we see, 442 flip-flops are clocked but only 441 receive a reset signal, this is due to the reset synchronization register being connected to the external reset signal rather than the internal reset tree. As the reset synchronization flip-flop is also not on the scan chain and we use full scan otherwise the 441 flip-flops on the scan tree match perfectly. You get the idea... Student Task 29: • Open the file chip.cts trace and search for Clock Tree to examine the leaf pins. • If everything looks OK we can proceed with clock synthesis. In the S YNTHESIZE C LOCK T REE form press OK. After a few minutes clock tree synthesis will be completed. Detailed reports will be generated under the directory specified on the form (most likely clock report). This directory includes a simple report file (clock.report).

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A summary report is also displayed on the C ADENCE S O C E NCOUNTER console. The first column shows the achieved performance while the second column reports the target specified in the configuration file. Student Task 30: • Check your results (summary and detailed reports). How many buffers were added? How many levels created? What’s the insertion delay? Are all constraints met? Note 1: You will get a max transition time violation on ClkxCI_PAD/I which can safely be ignored. As we have specified an input transition time of 800 ps on all primary inputs there is no way CTS could fulfill the 600 ps requirement at this point. Note 2: Unless the ‘‘RouteClkNet YES’’ option was used (more on this later), the timing figures reported are only estimates and might change quite a bit with detailed routing.

9 Timing Revisited At this point we will have to go into some more detail about timing. During different stages of the design flow, we have slightly different timing constraints (Refer to the following figure for the differences in the three stages). a) synthesis initially the design does not contain any pads. The input delay tidel and the output delay todel should contain the contribution of the input tinpad and output toutpad pads. b) pre-CTS during placement and routing phase, all required I/O pads and drivers will be present. At this stage there is no clock tree present. The timing should be adjusted, as at this moment the input delay tidel and output delay todel no longer include the pad delays. c) post-CTS once the clock tree is inserted, the timing will change slightly again. Due to the clock insertion delay tdi the internal clock will be slightly offset when compared to the external clock. At the input, the data travelling towards the first flip-flop inside the chip, will have more time, since this flip-flop will be trigerred by a clock signal that has been delayed by tdi . At the output however, the data that is coming from the chip will be launched with the internal clock, but will have to be sampled by the external clock. Consequently there will be less time for this signal. It should now be clear why it might be desirable to set constraints on the clock insertion delay property by specifying minimum and maximum values in the chip.ctstch file by MinDelay and MaxDelay parameters. The clock insertion delay can play an important part in the I/O delay. You may want to keep the insertion delay within certain limits to ensure proper I/O timing. Design tools have different mechanisms to deal with these three different cases. The simple solution is to use multiple constraint files for different stages. However, both S YNOPSYS D ESIGN C OMPILER and C ADENCE S O C E NCOUNTER accept several parameters to deal with this problem automatically. In the following we will discuss on how C ADENCE S O C E NCOUNTER calculates delays in the presence and absence of clock tree. The following table summarizes the most important settings:

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timing analysis mode

clock propagation mode

clock latency

(setAnalysisMode)

(set propagated clock)

(set clock latency)

-noSkew

forced ideal

no effect

-skew -noClockTree

forced ideal

SDCs in effect

-skew -clockTree a b

SDCs in

effecta

SDCs in effectb

still ideal mode unless set propagated clock is set set clock latency command is overridden by overlapping set propagated clock constraints

The timing analysis mode is automatically updated by C ADENCE S O C E NCOUNTER to match the design stage, i.e. before clock tree insertion it is set to ’-skew -noClockTree’ and afterwards to ’-skew -ClockTree’. The analysis mode can also be changed manually with the setAnalysisMode command. The two synopsys design constraints (SDC) set_propagated_clock and set_clock_latency are usually specified by the designer in the chip.sdc file. Furthermore, CTS tries to add a set_propagated_clock constraint on-the-fly (in memory), which can cause a number of problems: • This constraint will only be added if the AutoCTSRootPin pin/port in chip.ctstch and the clock waveform source pin/port (from the create_clock command in chip.sdc) are perfectly identical, i.e. not port vs. instance pin etc. • This constraint is never written to your chip.sdc file, so if you reload that file the constraint is lost. • Before CTS, only a pointer to your constraints file is saved along with the database. Now, if a constraint was added by CTS, all loaded constraints (including the new one) will be saved along with the database to a new file (*.pt). Restoring this database will then load this new constraints file instead of the one in encounter/src/ that you might have expected. Note: As soon as you manually (re-)load a constraints file, the behavior is reverted to the normal one. Now, as can be seen from the table above, to get the actual timing of the buffers/inverters on the clock tree instead of ideal mode, setting both ’-skew -ClockTree’ and set_propagated_clock is required. Also note that set_propagated_clock gets overridden for all pre-CTS design stages and could therefore be set right from the start (as already mentioned earlier). In ideal mode, the clock tree insertion delay is zero unless the set_clock_latency command is used to specify a different number, preferably close to the delay of the real tree (that is still to be inserted). While this ”placeholder” delay has the advantage that the I/O timing doesn’t change between pre-CTS and post-CTS phases, it renders timing reports more intransparent and is not handled exactly the same across different tools. Therefore, do not use this command unless you know what you are doing. In conclusion, it is recommended to include set_propagated_clock right from the start, not use set_clock_latency and load modified timing constraints after CTS only if required, i.e. when the I/O timing numbers (set_input_delay, set_output_delay) need to be adjusted to account for the actual clock tree29 . For this training we will modify and reload the constraints30 . 29 30

For slower clock speeds and/or uncritical I/O timing this is often not required. It might be more convenient to keep a separate post-CTS constraint file rather than changing the numbers back and fourth when redoing the flow.

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Tclk tidel tpd ff tpd a

tinpad

tpd b tsu ff tpd ff

a

a)

Tclk treg2reg

tin2reg

tpd c

b

c

tsu ff tpd ff tpd d

toutpad

tpd e tsu ff

d

e

Clk

tidel tpd ff tpd a

Tclk tin2reg

tinpad

Tclk

treg2reg

tpd b tsu ff tpd ff

a

tpd c

b

c

treg2out tsu ff tpd ff tpd d

todel toutpad

tpd e tsu ff

d

e

Chip Clk

Tclk tidel tpd ff tpd a

Tclk treg2reg

tin2reg tinpad

a

c)

todel

Top

Tclk

b)

Tclk treg2out

tpd b tsu ff tpd ff

tpd c

b

c

Tclk treg2out tsu ff tpd ff tpd d

todel toutpad

tpd e tsu ff

d

e

Chip Clk

tdi External Clock Internal Clock

Clock insertion delay More time for input

tidel

Less time for output

tin2reg

treg2out

todel

The previous figure illustrates all three stages in some detail. Whereever possible the same naming conventions as the textbook have been used31

31

Refer to page 235 “How to formulate timing constraints”, and page 346 “How to achieve friendly input/output timing” for more on this topic

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Student Task 31: • Copy your timing constraints file to filter_chip_postCTS.sdc and then modify the I/O timing constraints to account for the insertion delay of the actual clock tree, make sure that the clock is set to PROPAGATED MODE and load the constraints (Timing →Load \ Timing Constraint ...a ) • Run timing analysis (make sure to select P OST-CTS as design stage). • Examine the reports timingReports/chip postCTS*. You should now see the real timing on the clock network. • If you have violations, run a P OST-CTS (!) optimization with default settings. This should fix all violations. • Save the entire design. a

Currently loaded constraints will be purged before the new ones get loaded.

10 Signal Routing We will now route the signal nets. What you have seen so far are only trial-route nets that are not DRC clean and can therefore not be manufactured. Student Task 32: • There are two routing engines in C ADENCE S O C E NCOUNTER . WRoute is the older one and NanoRoute is supposed to be the latest and greatest. Start NanoRoute by selecting Route →NanoRoute →Route.... A large window will open. Enable the I NSERT D IODES option (you can leave the D IODE C ELL N AME field blank) and leave all other settings at their defaultsa . Click OK to start routing. You can observe the progress in the console window. a

On multi-CPU or multi-core machines you can increase the number of CPUs used by selecting Set Multiple CPU. This gives almost a linear speedup.

The F IX A NTENNA and I NSERT D IODE will cause the router to change layers and/or insert special protection diodes in order to avoid damages that can happen during manufacturing due to charges that accumulate on the wires and stress the gate oxide of input pins. Note that this is usually referred to as P ROCESS A NTENNAS which is entirely different from geometrical antennas (which is related to dangling wires).

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Our example design should route without problems. This is not always the case and we might get geometry violations. Geometry violations include shorts between nets and design rule violations (for example metal lines are drawn too close to be manufactured as separate wires). Needless to say that we must solve all these violations. You should always closely examine the violations in order to find out what causes them. Sometimes there is an unfortunate placement of macro-cells or power lines to blame and sometimes there is just not enough space to route all connections. Solutions range from re-running routing to completely reworking the floorplan. Student Task 33: • Now that we have the real signal wiring we need to perform a postroute timing analysis to see if we still meet all constraints. At this point not only a setup time analysis, but also a hold time analysis needs to be run. Usually it is not necessary to deal with hold time until this point. Note that you have to do two separate runs, one for setup and one for hold, as it is not possible do this in one single step. Use the GUI (make sure to select P OST-R OUTE) or type the commands below to perform the two analyses. enc > timeDesign -postRoute enc > timeDesign -postRoute -hold • Inspect the two summaries and the report files written to the ’timingReports’ directory. You will most likely have setup violations. To fix violations or increase the hold margin we can now perform a postroute optimization. Internal hold time violations need to be fixed in any case as, unlike internal setup violations, they can not be

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avoided later on (i.e. real chip) by lowering the clock speed32 . Further possibilities to improve timing include over-constraining the P OST-CTS optimization and enabling the T IMING D RIVEN option of NanoRoute. Earlier in the flow, T IMING D RIVEN P LACEMENT might be worth a try. Please note that the biggest improvements are possible with ’Pre-CTS’ optimization as the registers can be moved and resized at that stage. Per default, clock tree insertion will ”fix” the registers to preserve the clock tree, i.e. they no longer can be moved or resized. Student Task 34: • If you have large ”reg2reg” setup violations, this step may take a very long time. During the initial iterations of the design, it might be a good idea to use a more conservative (using a longer clock period) timing constraint so that not much time is spent during the optimization. Once you are satisfied with all other aspects of the design, you could revert to the original time constraints and let the optimizer try to achieve the timing. • Perform a postroute optimization Timing →Optimize .... • Optimization will delete and re-route all nets that are affected by the changes and run setup and hold mode timing analyses at the very end. Once again, inspect the reports.

Student Task 35: • Now let us have a look at the postroute timing of our clock tree(s) enc > reportClockTree -postRoute This will print a summary on the console and write a couple of report files chip.ctsrpt* to the encounter directory. There should be no (or only minor) violations of our clock tree constraints. Please note that the previous postCTS and postRoute setup (and hold) analyses already consider clock skew as they time every single path from the clock root to the leaf pins separately. Therefore, even a rather big skew reported here doesn’t really matter as long as the former analyses passed. So far, the clock tree has been routed as any other signal net. This is usually good enough, but if you want, for whatever reason, to further improve clock net timings, you can do the following (in CTS): 32

This does not necessarily hold true for multi-clock designs.

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• In the clock tree constraint file, set RouteClkNet YES. This is a per-tree setting that instructs CTS to call NanoRoute in order to route this clock net during clock tree insertion. The wires get a status of FIXED and will therefore not be changed later during signal routing. While this improves timing on the clock tree, overall routability gets worse. • To further improve timing, you can tell NanoRoute to route this net not like an ordinary signal net, but to create a balanced routing (by following the so called RouteGuide computed by CTS). To do so, set UseCTSRouteGuide YES in the clock constraint file33 .

11 Timing Debug To analyze timing violations, C ADENCE S O C E NCOUNTER also offers a graphical interface (Timing →Debug Timing) that visualizes paths and allows cross-probing with the layout. We will not explain the tool in detail here, but rather make some important notes: • This functionality is sort of standalone, it does not use results from the timeDesign command but runs a new analysis that generates the file top.mtarpt. Then these paths are visualized. • If the above file already exists, it will usually simply be loaded. This means that whenever your design has changed you have to regenerate this file in order to get up to date data. This can be done with the G ENERATE switch on the form that opens when you click the folder icon. • When generating the top.mtarpt, the current timing mode is relevant, i.e. to analyze hold paths timing mode has to be set to hold mode.

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This will persistently(!) alter the global CTS Mode to ‘‘setCTSMode -useCTSRouteGuide’’

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12 Finishing We are almost done with backend design, there are only a few steps required to finish the layout and verify that everything is correct.

12.1 Insert Filler Cells Student Task 36: • Now that we don’t need the additional space within the standard cell rows anymore, we have to fill these gaps with filler cells. This is required for fabrication. In addition, some of them contain capacitors between VCC and GND that filter spikes on the power lines. enc > source scripts/fillcore.tcl

Note that your row utilization will be 100% after this step. This means that you will have no room for further optimizations. Make sure to insert filler cells after all optimizations have been completed.

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Note: It is also possible to remove the filler cells with Place →Filler →Delete... or by using the script removefillcore.tcl.

12.2 Checking Connectivity and Geometry Violations Now that we are completely finished with the layout, we should make sure that we have no connection errors, i.e. all logic connections from the netlist are also present in the physical layout. Student Task 37: • Select Verify →Verify Connectivity ... from the menu. A window will appear. Run the analysis and check the console for the report summary. There should be no violations. • In a similar way let us verify all geometrical shapes. Select Verify →Verify Geometry \ ... from the menu. Run the analysis and check the report on the console. You should get no violations.

There is a script that will perform the last verification steps for you automatically. You can set a variable DESIGNNAME to assign the base name for all the files generated by this script. enc > set DESIGNNAME MyBeautifulChip enc > source scripts/checkdesign.tcl

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12.3 Evaluate the Physical Design Take the time to examine the routing. This is the main feedback you need for a second back-end iteration. Try to view all metal lines separately to see how congested your routing is. If you see a lot of Metal-6 (orange) you are probably close to the density limit. In our design you should not notice any congestion and Metal-6 will barely be used. If your design routed without problems and the routing was rather sparse then the next time you could assign a smaller core area and increase the row utilization. On the other hand if the design barely routed you have found the limits, in a second iteration you might consider assigning a little more core area timing degrades with congestion. Check the connections of your macro-cells and pads, this may give you an idea how to place the macro-cells the next time around. You need to get used to evaluating the result of different back-end design runs.

12.4 Generate Output Files Congratulations, you have completed the back-end design. That was not so hard now, was it? Student Task 38: • Save your design using Design →Save Design As ... →SoCE to the save directory and make sure that you use a name that shows this is a finished design (i.e. chip final.enc). • Finally we need to export all data needed for post layout simulation and physical verification (DRC/LVS). There is a script that will write out all relevant files to the out/ directorya . enc > source scripts/exportall.tcl a

To get complete supply net connectivity in the Verilog netlist for LVS, the missing connections for the power and ground pins (GNDIO/VCC3IO) of the pads are added and removed on-the-fly. We could also define and handle these two nets in the same way as VCC/GND, but there are more drawbacks than benefits.

Similar to the checkdesign.tcl file, the variable DESIGNNAME will be used to assign the base name of the files. If you do not specify a name, final will be used. After you complete this step you will have the following files: *.v This is the final netlist. Make sure to use this netlist for post layout simulations. *.gds.gz The layout in GDSII (Graphic Design System II) format. This is the standard format for exchanging layout data. *.sdf.gz The SDF (Standard Delay Format) file to be used for post layout simulation.

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