Topology Investigation for Front End DC-DC Power Conversion for Distributed Power System

Topology Investigation for Front End DC/DC Power Conversion for Distributed Power System

Bo Yang Dissertation submitted to the Faculty of the Virginia Polytechnic Institute and State University in partial fulfillment of the requirements for the degree of Doctor of Philosophy in Electrical Engineering

Fred C. Lee, Chairman

Dushan Boroyevich

Jason Lai

Guo-Quan. Lu

Alex Q. Huang September 12, 2003 Blacksburg, Virginia

Keywords: distributed power system, DC/DC converter, integrated magnetic, small signal model, resonant converter Copyright 2003, Bo Yang

Topology Investigation for Front End DC/DC Power Conversion for Distributed Power System by Bo Yang Fred C. Lee, Chairman Electrical engineering

(Abstract) With the fast advance in VLSI technology, smaller, more powerful digital system is available. It requires power supply with higher power density, lower profile and higher efficiency. PWM topologies have been widely used for this application. Unfortunately, hold up time requirement put huge penalties on the performance of these topologies. Also, high switching loss limited the power density achievable for these topologies. Two techniques to deal with hold up time issue are discussed in this dissertation: range winding solution and asymmetric winding solution, the efficiency at normal operation point could be improved with these methods. To reduce secondary rectifier conduction loss, QSW synchronous rectifier is developed, which also helps to achieve ZVS for symmetrical half bridge converter. Although with these methods, the efficiency of front end DC/DC converter could be improved, the excessive switching loss prohibited higher switching frequency. To achieve the targets, topologies with high switching frequency and high efficiency must be developed. Three resonant topologies: SRC, PRC and SPRC, are been investigated for this application because of their fame of low switching loss. Unfortunately, to design with hold up requirement, none of them could provide significant improvements over PWM converter.

Although the negative outcome, the desired characteristic for front end application could be derived. Base on the desired characteristic, a thorough search is performed for three elements resonant tanks. LLC resonant topology is found to posses the desired characteristic. From comparison, LLC resonant converter could reduce the total loss by 40% at same switching frequency. With doubled switching frequency, efficiency of LLC resonant converter is still far better than PWM converters. To design the power stage of LLC resonant converter, DC analysis is performed

with

two

methods:

simulation

and

fundamental component

simplification. Magnetic design is also discussed. The proposed integrated magnetic structure could achieve smaller volume, higher efficiency and easy manufacture. To make practical use of the topology, over load protection is a critical issue. Three methods to limit the stress under over load situation are discussed. With these methods, the converter could not only survive the over load condition, but also operate for long time under over load condition. Next small signal characteristic of the converter is investigated in order to design the feedback control. For resonant converter, state space average method is no longer valid. Two methods are used to investigate the small signal characteristic of LLC resonant converter: simulation and extended describing function method. Compare with test results, both methods could provide satisfactory results. To achieve both breadth and depth, two methods are both used to reveal the myth. With this information, compensator for feedback control could be designed. Test circuit of LLC resonant converter was developed for front end DC/DC application. With LLC topology, power density of 40W/in3 could be achieved compare with 15W/in3 for PWM converter.

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To My Wife Qiaoqiao

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Acknowledgement

ACKNOWLEDGMENT

I would like to thank my advisor, Dr. Fred C. Lee. Till today, I am still amused by his great intuition, broad knowledge and accurate judgment. The most precious thing I learned from him is the attitude toward research, which can be applied to every aspects of life too. Without his guidance and challenging, I will never be able to achieve this. I am grateful to my committee: Dr. Dushan Boroyevich, Dr. Jason Lai, Dr. Alex Q. Huang and Dr. Guo Q. Lu, and Dr. Dan Y. Chen for their valuable suggestions and numerous help. It has been a great pleasure to work in the Center for Power Electronics Systems (CPES). I would like to acknowledge the CPES administrative and management staff, Ms. Teresa Shaw, Ms. Linda Gallagher, Ms. Trish Rose, Ms. Ann Craig, Ms. Marianne Hawthome, Ms. Elizabeth Tranter, Ms. Lesli Farmer, Mr. Robert Martin, Mr. Steve Z. Chen, Mr. Dan Huff, Mr. Gary Kerr, and Mr. Jamie Evans for their countless help. I would like to thank my colleagues, Dr. Qiong Li, Dr. Xunwei Zhou, Dr. Pitleong Wong, Dr. Richard Zhang, Dr. Kun Xing, Dr. Jindong Zhang, Dr.Fengfeng Tao, Dr. Weiyun Chen, Mr. David Wen, Mr. Deng-Ming Peng, Mr. Zhou Chen, Mr. Yuanxuan Hu, Dr. Yong Li, Mr. Wenkang Huang, Dr. Sihua Wen, Mr. Rengang Chen, Miss Jinghong Guo, Mr. Wei Xu, Mr. Zhenxue Xu, Dr. v

Bo Yang

Acknowledgement

Ming, Xu, Mr. Ho-Pu Wu, Mr. Xiaowu Sun, Mr. Kaiwei Yao, Mr. Yuhui Chen, Mr. Mao Ye, Mr. Yu Meng, Mr. Yuancheng Ren, Prof. Wei Chen and Prof. Xiaochuan Jia. Their friendships and help have made my stay at CPES pleasant and enjoyable. I am especially indebted to my teammates in the DPS group, Dr. Peter Barbosa, Dr. Qun Zhao, Dr. Francisco Canales, Dr. Wei Dong, Mr. Bing Lu, Mr. Yang Qiu, Mr. Liyu Yang, Miss Manjing Xie, Miss Juanjuan Sun, Miss Tina Sang, and Mr. Dianbo Fu. It was a pleasure to work with such a talented, hardworking and creative group. My heartfelt appreciation goes toward my father, Huairui Yang, who has always encouraged me to pursue higher education. I also want thank my sister, Jing, for her love and support to me. With deepest love, I would like to thank my wife, Qiaoqiao, who has always been there with her love, support, understanding and encouragement for all my endeavors.

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Table of content

TABLE OF CONTENT Chapter 1 1.1 1.2 1.3

Chapter 2 2.1 2.2 2.3 2.4 2.5 2.6 2.7

Two and Three Components Resonant Tanks .................... 247

Two resonant components resonant tank............................................................................... 248 Three resonant components resonant tank............................................................................. 252

Appendix B B.1.

Summary and Future Work ..................................................... 237

Summary ...................................................................................................................................... 237 Future work................................................................................................................................. 243

Appendix A. A.1. A.2.

Small signal analysis and control design of LLC converter .. 187

Introduction................................................................................................................................. 187 Extended Describing Function analysis ................................................................................. 191 Time domain simulation method............................................................................................. 195 Impact of circuit parameters..................................................................................................... 218 Test verification.......................................................................................................................... 227 Compensator design for LLC resonant converter................................................................. 229 Summary ...................................................................................................................................... 236

Chapter 7 7.1 7.2

Improvements of LLC Resonant Converter ........................... 142

Magnetic design for LLC Resonant Converter..................................................................... 142 Over load protection for LLC resonant converter................................................................ 161 Integrated power electronics module for LLC ...................................................................... 180 Summary ...................................................................................................................................... 185

Chapter 6 6.1 6.2 6.3 6.4 6.5 6.6 6.7

LLC Resonant Converter............................................................ 94

Introduction................................................................................................................................... 94 Three traditional resonant topologies ....................................................................................... 95 LLC resonant converter............................................................................................................. 106 Operation of LLC resonant converter..................................................................................... 109 Power stage parameter design of LLC resonant converter................................................. 117 Extension of LLC resonant topology...................................................................................... 133 High frequency operation......................................................................................................... 135 Summary ...................................................................................................................................... 140

Chapter 5 5.1 5.2 5.3 5.4

Integrated Power Electronics Module ....................................... 72

Introduction................................................................................................................................... 72 Integrated Power Electronics Module for Front end DC/DC............................................... 75 Performance evaluation............................................................................................................... 85 Summary ........................................................................................................................................ 92

Chapter 4 4.1 4.2 4.3 4.4 4.5 4.6 4.7 4.8

State of the art topologies and improvements ........................... 18

Introduction................................................................................................................................... 18 State of the art topologies and issues ........................................................................................ 19 Issue with hold up time requirement......................................................................................... 28 Range winding for wide input range......................................................................................... 36 Asymmetrical Winding Asymmetrical Half Bridge............................................................... 44 QSW Synchronous Rectification............................................................................................... 54 Summary and issues .................................................................................................................... 67

Chapter 3 3.1 3.2 3.3 3.4

Introduction.................................................................................... 1

Background and Objectives.......................................................................................................... 1 State of the art topologies ............................................................................................................. 6 Outline of dissertation................................................................................................................. 11

Operation modes and DC analysis of LLC resonant converter 269

Operating modes of LLC resonant converter in Region 1 .................................................. 270

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B.2. B.3. B.4.

Table of content

Operating modes of LLC resonant converter in Region 2 .................................................. 276 Operating modes of LLC resonant converter in Region 3 .................................................. 282 DC analysis of LLC resonant converter................................................................................. 284

Appendix C. Appendix D. D.1. D.2.

Small signal characteristic of SRC converter...................... 289 LLC converter model for EDF analysis ............................... 293

Process of building LLC circuit model for EDF analysis ................................................... 293 LLC resonant converter model ................................................................................................ 304

Reference ...................................................................................................... 309 Vita ..................................................................................................................... 317

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List of illustrations

LIST OF ILLUSTRATIONS Figure 1.1. Distributed power system structure.......................................................................................... 2 Figure 1.2. Trend for server system power level........................................................................................ 4 Figure 1.3. Trend for AC/DC power supply for server............................................................................. 5 Figure 1.4. Trend toward lower profile system .......................................................................................... 5 Figure 1.5. Two stage structure of front-end converter............................................................................. 6 Figure 1.6 Primary inverter topologies ........................................................................................................ 7 Figure 1.7 Secondary rectifier topologies.................................................................................................... 7 Figure 1.8 The state of the art front-end system......................................................................................... 9 Figure 2.1. Two-switch forward converter and operating waveforms .................................................. 20 Figure 2.2. Full bridge converter and operating waveforms .................................................................. 22 Figure 2.3. Schematic of half bridge converter........................................................................................ 24 Figure 2.4. Operating waveforms of symmetrical half bridge converter............................................. 24 Figure 2.5. Operating waveforms of asymmetrical half bridge converter........................................... 25 Figure 2.6. Asymmetrical half bridge with current doubler and waveforms ....................................... 26 Figure 2.7. Loss comparison of three state of the art topologies........................................................... 27 Figure 2.8. Output filter inductor volt-sec comparison of three topologies......................................... 28 Figure 2.9. Capacitance for hold up time and DC bus voltage after hold up time ............................. 30 Figure 2.10. Duty cycle range for asymmetrical half bridge with hold up requirement.................... 31 Figure 2.11. Secondary rectifier voltage and current stress for asymmetrical half bridge................ 32 Figure 2.12. Test efficiency of asymmetrical half bridge....................................................................... 33 Figure 2.13. Duty cycle range for full bridge converter with hold up requirement ........................... 34 Figure 2.14. Circulating current of full bridge with different input voltage........................................ 35 Figure 2.15. Asymmetrical half bridge converter with range winding................................................. 37 Figure 2.16 Circuit diagram for normal operation ................................................................................... 38 Figure 2.17 Circuit diagram for hold up operation .................................................................................. 38 Figure 2.18 Duty cycle range comparison of asymmetrical half bridge with/without range switch .................................................................................................................................................................. 40 Figure 2.19 Diode voltage stress comparison........................................................................................... 41 Figure 2.20 Diode average current comparison........................................................................................ 41 Figure 2.21 Primary switch RMS current comparis on............................................................................ 42 Figure 2.22 Primary switch turn off current comparison........................................................................ 42 Figure 2.23 Test setup for range winding asymmetrical half bridge .................................................... 43 Figure 2.24 Test efficiency at normal operation for range winding solution...................................... 43 Figure 2.25 Range winding for (a) Full bridge (b) current doubler...................................................... 44 Figure 2.26 Half bridge current doubler with (a) one transformer and (b) Two transformers ......... 45 Figure 2.27 Topology transformation of current doubler....................................................................... 46 Figure 2.28 DC characteristic for different np ratio................................................................................. 47 Figure 2.29 Duty cycle range with different turn ratios.......................................................................... 48 Figure 2.30 Operation waveforms of (a) asymmetrical half bridge and (b) asymmetrical winding asymmetrical half bridge ..................................................................................................................... 50 Figure 2.31 Diode voltage stress for asymmetrical winding asymmetrical half bridge .................... 51 Figure 2.32 Diode current stress for asymmetrical winding asymmetrical half bridge..................... 51 Figure 2.33 Output current ripple for asymmetrical winding asymmetrical half bridge................... 52 Figure 2.34 Test circuit diagram of asymmetrical winding asymmetrical half bridge ...................... 53 Figure 2.35 Test waveforms of asymmetrical winding asymmetrical half bridge ............................. 53 Figure 2.36 Test efficiency of asymmetrical winding asymmetrical half bridge ............................... 53 Figure 2.37 Rectifier diode voltage stress of different topologies ........................................................ 55 Figure 2.38 Conduction loss comparison of different devices............................................................... 55 Figure 2.39 Rectifier diodes current stress for asymmetrical winding asymmetrical half bridge ... 57 Figure 2.40 Conduction loss comparison of 200V diode and MOSFET ............................................. 57

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Figure 2.41 Circuit diagram of half bridge current doubler with synchronous rectifier.................... 58 Figure 2.42 Test waveform of synchronous rectifier for front end converter' .................................... 58 Figure 2.43 Operation waveform of synchronous rectifier for front-end converter........................... 59 Figure 2.44 Operation waveform of QSW synchronous rectifier.......................................................... 60 Figure 2.45 Operating mode 1 of half bridge with synchronous rectifier............................................ 61 Figure 2.46 Operating mode 2 of half bridge with synchronous rectifier............................................ 61 Figure 2.47 Operating mode 3 of half bridge with synchronous rectifier............................................ 61 Figure 2.48 Operating mode 4 of half bridge with synchronous rectifier............................................ 62 Figure 2.49 Operating mode 5 of half bridge with synchronous rectifier............................................ 62 Figure 2.50 Primary loss comparison of QSW and conventional synchronous rectifier................... 64 Figure 2.51 Test circuit of QSW synchronous rectifier .......................................................................... 64 Figure 2.52 Test waveform of QSW synchronous rectifier at light load ............................................. 65 Figure 2.53 Test waveform of QSW synchronous rectifier at heavy load........................................... 65 Figure 2.54 Test efficiency of half bridge with QSW synchronous rectifier...................................... 66 Figure 2.55 Full bridge with QSW synchronous rectifier....................................................................... 66 Figure 2.56 Efficiency comparison of different methods....................................................................... 67 Figure 2.57. Primary switch total loss for different switching frequency............................................ 69 Figure 2.58 Half bridge converter with snubber circuit .......................................................................... 69 Figure 2.59 Test waveform with and without snubber circuit ............................................................... 70 Figure 2.60 Prototype of 200kHz asymmetrical half bridge with discrete components.................... 70 Figure 3.1 Totem pole switches with parasitic inductance and Q2 Vds (100V/div).......................... 73 Figure 3.2 Impact of gate loop parasitic .................................................................................................... 74 Figure 3.3 Schematic of front end DC/DC with asymmetrical half bridge current doubler............. 75 Figure 3.4 System partitioning of front end DC/DC converter.............................................................. 77 Figure 3.5 (a) D2BGA IGBT chip-scale package, (b) schematic of FCOF power switching stage module structure, and (c) FCOF power switching module prototype. ........................................ 78 Figure 3.6 (a) Schematic of Dimple Array Interconnect, (b) integrated DAI power switching stage module, and (c) prototype Dimple Array Interconnect power switching stage module........... 79 Figure 3.7 (a) Schematic integration structure of emb edded power module and (b) circuit diagram .................................................................................................................................................................. 81 Figure 3.8 Assembly process of embedded power module: (a) top view of embedded power stage, (b) back view of embedded power stage, (c) components attachment on top, (d) patterned DBC for base substrate, (e) soldered on substrate, and (f) final encapsulated module. ........... 82 Figure 3.9 (a) Spiral integrated LC structure with distributed capacitance and possible external connection configurations, (b) simplified equivalent circuit, and (c) exploded view............... 83 Figure 3.10 Magnetic integrated for asymmetrical half bridge converter............................................ 84 Figure 3.11 Planar integrated magnetic for current doubler rectifier.................................................... 84 Figure 3.12 Explored view and photo of passive IPEM ......................................................................... 85 Figure 3.13 Gate charge curve of IXFH21N50 (a) Datasheet, (b) simulated...................................... 86 Figure 3.14 V/I characteristic of IXFH21N50 (a) datasheet and (b) simulated.................................. 87 Figure 3.15 Parasitic of (a) discrete MOSFET and (b) Active IPEM ................................................... 87 Figure 3.16 Simulation waveforms of Q1 drain source voltage (a) Discrete, and (b) IPEM ........... 88 Figure 3.17 Active IPEM turn-off loss reduction..................................................................................... 89 Figure 3.18 Voltage overshoots improvement by IPEM ........................................................................ 89 Figure 3.19 Photos of three prototypes to be compared (a) discrete design, (b) integrated magnetic design and (c) Passive IPEM design.................................................................................................. 90 Figure 3.20 Test efficiency of three prototypes........................................................................................ 90 Figure 3.21 Temperature test setup for three prototypes ........................................................................ 91 Figure 4.1 Half Bridge Series Resonant Converter.................................................................................. 96 Figure 4.2 DC characteristic and operating region of SRC.................................................................... 97 Figure 4.3 Simulation waveforms of SRC................................................................................................. 97 Figure 4.4 Half bridge parallel resonant converter.................................................................................. 99 Figure 4.5 DC characteristic and operating region of PRC.................................................................. 100 x

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Figure 4.6 Simulation waveforms of PRC............................................................................................... 101 Figure 4.7 Half bridge series parallel resonant converter..................................................................... 103 Figure 4.8 DC characteristic and operating region of SPRC................................................................ 104 Figure 4.9 Simulation waveforms of SPRC............................................................................................ 104 Figure 4.10 LCC and LLC resonant tank................................................................................................ 107 Figure 4.11 DC characteristic of LCC resonant converter................................................................... 108 Figure 4.12 DC characteristic of LLC resonant converter.................................................................... 108 Figure 4.13 Half bridge LLC resonant converter................................................................................... 108 Figure 4.14 DC characteristic of LLC resonant converter.................................................................... 111 Figure 4.15 Three operating regions of LLC resonant converter........................................................ 111 Figure 4.16 Simulated operation waveforms in region 1 ...................................................................... 112 Figure 4.17 Simulation waveforms in region 2 ...................................................................................... 113 Figure 4.18 Circuit diagram during mode 1 in region 2........................................................................ 114 Figure 4.19 Circuit diagram during mode 2 in region 2........................................................................ 115 Figure 4.20 Circuit diagram during mode 3 in region 2........................................................................ 116 Figure 4.21 Operating region for design 1 .............................................................................................. 119 Figure 4.22 Simulation waveforms of design 1 with 300V and 400V input voltage....................... 119 Figure 4.23 Operating region for design 2 .............................................................................................. 120 Figure 4.24 Simulation waveforms of design 2 with 300V and 400V input voltage....................... 120 Figure 4.25 Operating region for design 3 .............................................................................................. 121 Figure 4.26 Simulation waveforms of design 3 with 300V and 400V input voltage....................... 121 Figure 4.27 Primary loss for three designs with Vin=400V................................................................. 122 Figure 4.28 Primary loss for three designs with Vin=360V................................................................. 123 Figure 4.29 Primary loss for three designs with Vin=300V................................................................. 123 Figure 4.30 Test circuit for 200kHz LLC resonant converter.............................................................. 125 Figure 4.31 Test waveform of LLC converter at full load and (a) 300V input, and (b) 400V input ................................................................................................................................................................ 126 Figure 4.32 Test efficiency of LLC converter and HB converter at 400V input.............................. 126 Figure 4.33 Test efficiency of LLC and AHB converter at different input voltage and full load . 127 Figure 4.34 Two PWM topologies to be compared (a) AHB, and (b) asymmetrical winding AHB ................................................................................................................................................................ 128 Figure 4.35 Primary side conduction loss comparison.......................................................................... 130 Figure 4.36 Primary switches turn off loss comparison........................................................................ 131 Figure 4.37 Secondary conduction loss comparison.............................................................................. 132 Figure 4.38 Resonant tanks with desired DC characteristic ................................................................. 134 Figure 4.39 Half bridge resonant converter with tank W...................................................................... 135 Figure 4.40 Half bridge resonant converter with tank U ...................................................................... 135 Figure 4.41 Simulation waveform of LLC resonant converter............................................................ 136 Figure 4.42 Switching loss with different Lm for 400kHz LLC resonant converter....................... 137 Figure 4.43 Circuit diagram of 400kHz LLC resonant converter....................................................... 137 Figure 4.44 Integrated magnetic structure for 400kHz LLC resonant converter.............................. 137 Figure 4.45 Magnetic components size comparison (a) 200kHz and (b) 400kHz............................ 138 Figure 4.46 Test circuits of front end DC/DC converter (a) 200kHz AHB, (b) 200kHz LLC, and (c) 400kHz LLC.................................................................................................................................. 139 Figure 4.47 Test efficiency of 400kHz LLC resonant converter......................................................... 140 Figure 5.1 Magnetic structure for LLC resonant converter.................................................................. 143 Figure 5.2 Discrete magnetic design (a) schematic (b) physical structure ........................................ 144 Figure 5.3 Flux density simulation result (a) Inductor, and (b) Transformer.................................... 144 Figure 5.4 Peak to peak flux density under different input voltage at full load................................ 145 Figure 5.5 Structure with real transformer .............................................................................................. 147 Figure 5.6 Desired magnetic components configuration ...................................................................... 147 Figure 5.7 Magnetic components configuration from real transformer............................................. 148 Figure 5.8 Voltage stress of output diodes D1 D2 (a): desired structure (b) real transformer ....... 148 xi

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Figure 5.9 Integrated Magnetic Designs A.............................................................................................. 149 Figure 5.10 Flux density simulation result for Design A...................................................................... 150 Figure 5.11 Center leg flux density for different input voltage........................................................... 150 Figure 5.12 general magnetic structures for Integrated magnetic ....................................................... 152 Figure 5.13 Reluctance model of general integrated magnetic structure ........................................... 153 Figure 5.14 Circuit model of general integrated magnetic structure .................................................. 153 Figure 5.15 Integrated Magnetic Designs B............................................................................................ 154 Figure 5.16 Electrical circuit model of integrated magnetic structure B ........................................... 154 Figure 5.17 Electrical model of connecting dot-marked terminal with unmarked terminal........... 155 Figure 5.18 Two operation modes for LLC resonant converter.......................................................... 155 Figure 5.19 Equivalent-circuit for mode (a)............................................................................................ 156 Figure 5.20 Design curves for integrated magnetic structure B for LLC converter......................... 158 Figure 5.21 Flux density in each leg for integrated magnetic structure B ......................................... 159 Figure 5.22 Efficiency comparison of integrated and discrete magnetic design for LLC converter ................................................................................................................................................................ 160 Figure 5.23 Magnetic size comparison of discrete and integrated magnetic ..................................... 160 Figure 5.24 Simulation waveforms of LLC resonant converter at (a) normal operation, and (b) short circuit operation ........................................................................................................................ 164 Figure 5.25 Lost of ZVS for LLC resonant converter during over load situation............................ 165 Figure 5.26 High current stress during over load situation for LLC resonant converter................. 165 Figure 5.27 Simplified model of LLC resonant converter during short circuit condition............... 166 Figure 5.28 Short circuit output current at different switching frequency......................................... 167 Figure 5.29 Average output current vs. switching frequency under short circuit............................. 168 Figure 5.30 Change of operating mode with different switching frequency under protection mode ................................................................................................................................................................ 168 Figure 5.31 Test waveform (top to bottom: Q1 gate signal, Transformer primary current and resonant cap voltage).......................................................................................................................... 169 Figure 5.32 Problems with high switching frequency protection mode............................................. 169 Figure 5.33 Control Method of Variable freq + PWM control............................................................ 171 Figure 5.34 Average output current of LLC converter with variable frequency + PWM control. 171 Figure 5.35 Simulation waveform for D=0.5, 0.4 and 0.2 at short circuit condition....................... 172 Figure 5.36 Simulation waveform with PWM control (from top: gate signal of Q1 and Q3, Vds of Q1 and primary current Ip)................................................................................................................ 172 Figure 5.37 Test waveform for PWM control (Top: Vds of Q1, middle: gate signal of Q1, and Q2, bottom: primary current) ................................................................................................................... 173 Figure 5.38 Two LLC resonant converter topologies: (a) Original LLC converter and (b) proposed clamped LLC converter..................................................................................................................... 174 Figure 5.39 Simulation waveforms for two LLC resonant converter topologies: (a) original LLC converter and (b) clamped LLC converter ..................................................................................... 175 Figure 5.40 Simulation waveforms under over load condition for (a) original LLC converter and (b) clamped LLC converter............................................................................................................... 176 Figure 5.41 State plane of original and clamped LLC resonant converter........................................ 177 Figure 5.42 Average output current under over load condition for original LLC converter.......... 177 Figure 5.43 Average output current under over load condition for clamped LLC converter......... 178 Figure 5.44 Design region of clamped LLC resonant converter ......................................................... 179 Figure 5.45 Test waveform of LLC converter under clamping mode ................................................ 179 Figure 5.46 LLC resonant converter with splitting resonant cap and clamping diodes .................. 181 Figure 5.47 Schematics of passive IPEM for AHB and LLC r converter.......................................... 182 Figure 5.48 Capacitor integration for LLC resonant converter........................................................... 182 Figure 5.49 Two passive IPEM structures for LLC resonatn covnerter............................................. 183 Figure 6.1 LLC resonant converter with feedback control................................................................... 188 Figure 6.2 Circuit parameters for extended describing function analysis .......................................... 192 Figure 6.3 Impact of harmonic order on the accuracy of EDF method in region 1.......................... 193 xii

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Figure 6.4 Impact of harmonic order on the accuracy of EDF method in region 2.......................... 194 Figure 6.5 Procedure for simulation method to analyze small signal characteristic ........................ 195 Figure 6.6 Circuit setup for first step simulation.................................................................................... 196 Figure 6.7 Circuit setup for second step simulation............................................................................... 196 Figure 6.8 System poles and zeros of LLC in region 1 with different switching frequency.......... 198 Figure 6.9 Input conductance of LLC converter in region 1 ................................................................ 199 Figure 6.10 Output impedance of LLC resonant converter in region 1.............................................. 199 Figure 6.11 Audio susceptibility of LLC converter in region 1 .......................................................... 200 Figure 6.12 System poles and zeros of LLC converter in region 2..................................................... 201 Figure 6.13 Input conductance of LLC resonant converter in region 2.............................................. 201 Figure 6.14 Output impedance of LLC resonant converter in region 2.............................................. 202 Figure 6.15 Audio susceptibility of LLC resonant converter in region 2 .......................................... 202 Figure 6.16 LLC converter setup for small signal analysis .................................................................. 204 Figure 6.17 Operating region of LLC resonant converter.................................................................... 205 Figure 6.18 Bode plot of control to output transfer function for LLC resonant converter.............. 205 Figure 6.19 Bode plot of control to output transfer function of LLC resonant converter in region 1 ................................................................................................................................................................ 206 Figure 6.20 Bode plot of control to output transfer function of LLC resonant converter in region 2 ................................................................................................................................................................ 207 Figure 6.21 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 1(fr=250kHz, fs=300kHz) ............................................................................... 210 Figure 6.22 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 1(fr=250kHz, fs=300kHz) (full load to 25% load) ..................................... 211 Figure 6.23 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 1(fr=250kHz, fs=300kHz) (25% to no load)................................................ 212 Figure 6.24 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) ............................................................................... 214 Figure 6.25 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) (full load to 25% load) ..................................... 215 Figure 6.26 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) (25% to 10% load)............................................ 216 Figure 6.27 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) (10% to no load)................................................ 217 Figure 6.28 Simulation setup for output capacitor impact on small signal characteristic ............... 218 Figure 6.29 Bode plot of control to output transfer function with different output capacitance with switching frequency 300kHz(region 1)........................................................................................... 219 Figure 6.30 Bode plot of control to output transfer function with different output capacitance with switching frequency 200kHz(region 2)........................................................................................... 220 Figure 6.31 Simulation setup for magnetizing inductance impact on small signal characteristic . 221 Figure 6.32 Bode plot of control to output transfer function with different magnetizing inductance with switching frequency 300kHz(region 1).................................................................................. 222 Figure 6.33 Bode plot of control to output transfer function with different magnetizing inductance with switching frequency 200kHz(region 2).................................................................................. 223 Figure 6.34 Simulation setup for resonant tank impedance impact on small signal characteristic 224 Figure 6.35 Bode plot of control to output transfer function with different resonant inductance with switching frequency 300kHz(region 1).................................................................................. 225 Figure 6.36 Bode plot of control to output transfer function with different resonant inductance with switching frequency 200kHz(region 2).................................................................................. 226 Figure 6.37 Test setup up for small signal characterization of LLC converter................................. 227 Figure 6.38 Bode plot of control to output transfer function at full load in region 1....................... 228 Figure 6.39 Bode plot of control to output transfer function at full load in region 2....................... 229 Figure 6.40 Compensator for PWM converter....................................................................................... 230 Figure 6.41 Compensator structures for LLC resonant converter....................................................... 230 xiii

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Figure 6.42 Small signal characteristic of LLC converter in region 2................................................ 231 Figure 6.43 Load impact on small signal characteristic of LLC converter in region 2 ................... 231 Figure 6.44 Small signal characteristic of LLC converter in region 1................................................ 232 Figure 6.45 Load impact on small signal characteristic of LLC converter in region 1 ................... 233 Figure 6.46 Compensator designed for LLC resonant converter......................................................... 234 Figure 6.47 Plant bode plot and loop gain bode plot in region 1......................................................... 234 Figure 6.48 Plant bode plot and loop gain bode plot in region 2......................................................... 235 Figure 6.49 Test result of load change from no load to full load ........................................................ 235 Figure 6.50 Test result of load change from full load to no load ........................................................ 235 Figure 7.1 LLC resonant tank and LLC resonant tank with passive current shaping...................... 244 Figure 7.2 Simulation waveform of LLC resonant converter.............................................................. 244 Figure 7.3 Simulation waveform of LLC resonant converter with passive current shaping.......... 244 Figure A.1 Input type for DC/DC converter........................................................................................... 248 Figure A.2 Input type for DC/DC converter........................................................................................... 248 Figure A.3 Two components resonant tanks ........................................................................................... 249 Figure A.4 DC characteristic of two components tank A ..................................................................... 250 Figure A.5 DC characteristic of two components tank B ..................................................................... 250 Figure A.6 DC characteristic of two components tank C ..................................................................... 251 Figure A.7 DC characteristic of two components tank D ..................................................................... 251 Figure A.8 Components configuration for three components resonant tank..................................... 252 Figure A.9. Resonant tank for components configuration 1................................................................. 253 Figure A.10. Resonant tank for components configuration 2 .............................................................. 253 Figure A.11. Resonant tank for components configuration 3 .............................................................. 253 Figure A.12. Resonant tank for components configuration 4 .............................................................. 254 Figure A.13. Resonant tank for components configuration 5 .............................................................. 254 Figure A.14. Resonant tank for components configuration 6 .............................................................. 254 Figure A.15. Resonant tank for components configuration 7 .............................................................. 255 Figure A.16. Resonant tank for components configuration 8 .............................................................. 255 Figure A.17. Resonant tank for components configuration 9 .............................................................. 255 Figure A.18. Resonant tank for components configuration 10............................................................ 256 Figure A.19. Resonant tank for components configuration 11............................................................ 256 Figure A.20. Resonant tank for components configuration 12............................................................ 257 Figure A.21. Resonant tank for components configuration 13............................................................ 257 Figure A.22. Resonant tank for components configuration 15............................................................ 258 Figure A.23. DC characteristic of tank C ................................................................................................ 260 Figure A.24. DC characteristic of tank D................................................................................................ 261 Figure A.25. DC characteristic of tank G................................................................................................ 261 Figure A.26. DC characteristic of tank H................................................................................................ 262 Figure A.27. DC characteristic of tank K................................................................................................ 262 Figure A.28. DC characteristic of tank L................................................................................................. 263 Figure A.29. DC characteristic of tank O................................................................................................ 263 Figure A.30. DC characteristic of tank S................................................................................................. 264 Figure A.31. DC characteristic of tank U................................................................................................ 264 Figure A.32. DC characteristic of tank V................................................................................................ 265 Figure A.33. DC characteristic of tank W ............................................................................................... 265 Figure A.34. DC characteristic of tank X................................................................................................ 266 Figure A.35. DC characteristic of tank Z................................................................................................. 266 Figure A.36. DC characteristic of tank G1 .............................................................................................. 267 Figure A.37. DC characteristic of tank H1.............................................................................................. 267 Figure B.1 DC characteristic of LLC resonant converter..................................................................... 270 Figure B.2 Waveform of operation mode 1 in region 1 for LLC resonant converter' ..................... 271 Figure B.3 Waveform of operation mode 2 in region 1 for LLC resonant converter' ..................... 273 Figure B.4 Waveform of operation mode 3 in region 1 for LLC resonant converter' ..................... 275 xiv

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Figure Figure Figure Figure Figure Figure Figure Figure Figure Figure Figure Figure Figure Figure Figure

List of illustrations

B.5 Waveform of operation mode 1 in region 2 for LLC resonant converter' ..................... 277 B.6 Waveform of operation mode 2 in region 2 for LLC resonant converter' ..................... 279 B.7 Waveform of operation mode 3 in region 2 for LLC resonant converter' ..................... 281 B.8 Waveform in region 3 for LLC resonant converter........................................................... 283 B.9 Simplified topology with fundamental component assumption ...................................... 284 B.10 DC characteristic from simplified model .......................................................................... 286 B.11 DC characteristic from simulation method....................................................................... 286 B.12 Error of simplified circuit model ........................................................................................ 287 C.1 SRC circuit for small signal analysis ................................................................................... 289 C.2 Bode plot of control to output transfer function of Series Resonant Converter............ 290 D.1 Circuit diagram and notification for extended describing function analysis................. 294 D.2 Simulation waveform of LLC converter in region 1 ......................................................... 295 D.3 Topology modes and progressing condition for region 1 ................................................. 296 D.4 Simulation waveform of LLC converter in region 2 ......................................................... 300 D.5 Topology modes and progressing condition for region 2 ................................................. 300

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List of Table

LIST OF TABLES Table 2-1. Input range of front-end DC/DC converter vs. DC bus capacitance................................. 30 Table 2-2 Difference between two asymmetrical half bridge design ................................................... 33 Table 3-1 Comparison of three prototypes ................................................................................................ 91 Table 3-2 Temperature test results of three prototypes ........................................................................... 91 Table 4-1 Summary of three LLC resonant converter designs ............................................................ 122 Table 4-2 Loss breakdown comparison of AHB and LLC................................................................... 132 Table A-1 Classification of three components resonant tanks ............................................................. 259 Table A-2 Three components resonant tanks with voltage source input............................................ 260

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Chapter 1. Introduction

1.1 Background and Objectives With fast advance in very large-scale integrated circuit (VLSI) technology, more and more transistors can be integrated into smaller silicon chips. As a result, more powerful, compact digital systems are becoming available. At the same time, these exciting changes in VLSI also imposed exciting challenges on power management for these digital systems. The challenges come from several aspects of changes in digital system. First, as more and more transistors are integrated into the integrated circuit chip, the power required to operate the chip is increasing very rapid. Second, with the transistors working at higher frequency, the supply voltage is reducing with fast transient speed and tight regulation requirement. Third, as VLSI technology is moving very fast, the power management requirement is becoming a fast moving target. Distribute power system (DPS) as shown in Figure 1.1, is been widely used for server and telecom power systems which represents the most advanced digital systems. In a distributed power system, power is processed by two stages. First stage converts AC input to 48V intermediate DC bus. This DC voltage is then distributed to the load side. The load converter, which is located on the load side,

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processes the power second time by converting DC distribution bus to whatever voltage needed by the load.

Figure 1.1. Distributed power system structure

Many advantages of distributed power system prompted its use in these applications. First, with fast dropping on supply voltage of digital system, it is not realistic to delivery the power with such low voltage. DPS uses much high voltage to distribute power. This greatly reduces the loss associated with power distribution. Second, since the second stage (load converter) is placed very close to the load. The impact of parasitic is minimized. This converter can have very fast transient response to provide the fast current slew rate to the load. Third, for a distributed power system, front-end converter is independent of the load requirement. Each load converter is also independent to other load. This provides significant benefit for the fast changing system requirement. With distributed power system, when technology changes, only the load converter associated with that load need to be redesigned, the impact on whole system is minimized. 2

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Beside these aspects, DPS also provides other benefits. First, DPS is an open architecture, modularized solution. The power system can be reconfigured as load system been expended or upgraded. It is a system which can grow as needed. Second, with modularized design, high reliability can be achieved with N+1 redundancy [A7]. Because of these advantages of DPS, it is been widely adopted for different applications [A4][A5][A6][A8]. DPS is been adopted exclusively in high end sever system and telecommunication system. Even for the most cost sensitive application like Personal Computer, DPS concept is been partly adopted. For today's personal computer, a hybrid power system is used. For critical components like CPU and Video Adapter, the distributed power system concept is used. For less critical components like modem-card or network-card, it still use centralized power system structure. With increased clock speed, very soon the memory will also have dedicated power supply. More DPS structure could be expected. Although for DPS, the front-end converter is not so closely related to the load, still other aspects of the load requirement imposes lot of new challenges to the front-end converter. The major impacts come from following aspects. First, as integration level increases with surprising speed, more and more transistors are integrated to the system with faster switching frequency. The digital system is becoming more and more power hungry and compact. As a supporting subsystem,

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people tend to give fewer budgets for the power supply system. So the requirement for the power system is to provide higher power with smaller volume. Another significant difference, which is driving the industry, is the profile. For the digital system, all the components now can be build with very low profile. So people expect to build the system with low profile too. With lower profile, more computational power could be build into smaller rack; this will reduce the system and maintains cost. This calls for a power supply to have compatible profile with digital components. Traditional systems normally have a profile of 1.5U to 2U (1U=1.75inch), now the industry is moving toward 1U power system. These trends could be observed in Figure 1.2, Figure 1.3 and Figure 1.4. In Figure 1.2 [A1], the trend for computer server system power requirement is shown. For high-end server system, the power level will increase by factor of 6 in recent 5 years.

(From IBM Power Technology Symposium 2002, by Dr. Thai Q. Ngo, IBM) Figure 1.2. Trend for server system power level

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In Figure 1.3[A1], the trends for power density, efficiency and lifetime are shown. Within next several years, the power density needs to increase by a factor of 2. Efficiency needs to increase by more than 5%. To achieve this efficiency improvement, 30 to 50% reduction of system loss is required.

(From IBM Power Technology Symposium 2002, by Dr. Thai Q. Ngo, IBM) Figure 1.3. Trend for AC/DC power supply for server

In Figure 1.4[A1], the need for 1U system is showed for 2000 and 2001. As seen in the picture, within one years time, the need for 1U system doubled. As for now, more and more server system are built with 1U profile as can be seen in all the major server manufactures like DELL, APPLE, GATEWAY etc.

(From IBM Power Technology Symposium 2002, by Anthony Stratakos, Volterra) Figure 1.4. Trend toward lower profile system

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From above discussions, we can see that the trends for front-end system are strongly affected by the digital system evolutions. The power density is expected to double. Loss is expected to reduce by more than 30%. And profile is expected to reduce by 50%. Next the state of the art technology will be reviewed and paths to achieve these goals will be discussed.

1.2 State of the art topologies Inside front-end converter, two-stage approach is widely adopted as shown in Figure 1.5. With two-stage approach, there are two power conversion stages inside the front-end converter. First stage converts AC input to a loosely regulated 400V intermediate DC bus with power factor correction. Second stage, front-end DC/DC converter, will convert 400V DC into a tightly regulated 48V DC bus, which will be distributed to the load converter. For a single-phase system, 1kW system is the most popular power level because of its flexibility for 2 to 3 kW system. Also, at this power level, the choice of power devices is around the optimal.

Figure 1.5. Two stage structure of front-end converter

In this dissertation, the design challenges, issues and solutions for the second stage – Front-end DC/DC converter in a 1kW system will be discussed.

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Most front-end DC/DC converter designs evolve around full-bridge, twoswitch forward, and half-bridge converters, as shown in Figure 1.6. Among the possibilities and for the power level under consideration (1 kW), half-bridge converter and full bridge converter provide the best combination of simple structure, low device stress and soft switching capability. Most industry products use these two topologies.

Figure 1.6 Primary inverter topologies

Figure 1.7 Secondary rectifier topologies

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Figure 1.7 shows possible configurations for secondary-side rectification. It is well understood that certain combinations of primary- and secondary-side topologies are deemed less desirable. Each secondary-side rectification has its advantages and disadvantages, which will be discussed in detail in chapter 2. In Figure 1.8, a state of the art front-end system use full bridge and center tapped rectifier is shown. The magnetic components and heat sinks in the system are outlined. The upper half part of the picture is dedicated to PFC converter and lower part is the front end DC/DC converter. For DC/DC part, it is clearly that heat sink and magnetic are the biggest parts, which occupied more than 80% of the total system volume. To improve power density and profile, size reduction of heat sink and magnetic components are necessary. Several methods could be used to reduce the heat sink and magnetic: High switching frequency: higher switching frequency could result in volume reduction of passive components. High efficiency: thermal management is a big part of the system. To achieve high power density, reduction on the volume for thermal management is an effective way.

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Figure 1.8 The state of the art front-end system

However, these two methods don’t come together easily. With high switching frequency, efficiency often will suffer. The reduced efficiency is because of high switching loss and also the reverse recovery of the secondary-side diodes. The diode re-verse recovery causes voltage overshoot and ringing across the devices, which impacts on the selection of device breakdown voltage. The loss due to diode reverse recovery is also a great part of the total loss. Snubbers (such as those with active clamping or saturable cores) are used to deal with this problem. Nevertheless, these solutions also have limitations. Due to the large current in the secondary side, the conduction loss of the diode is another important part of the overall loss. The use of Schottky diodes can reduce the reverse-recovery problem. For the front-end dc–dc converter, the secondary diode voltage stress is normally close to 200 V. As a result, it is difficult to find suitable Schottky diodes for such a high voltage, and then other solutions are required.

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Other than high switching loss, several other obstacles prevent us from switching so fast. High stress and high EMI noise caused by parasitic components are two major problems too. Another obstacle for front end DC/DC converter design is the hold up time requirement: when the input AC line is gone, system needs to work for 20ms with full power. With hold up time, the result is wide input range for front end DC/DC converter. The performance at high input voltage is critical to system power density and efficiency while performance at low input voltage is not so important. For current state of the art topologies, however, this wide input range greatly impairs the performance of the converter at high input voltage. To overcome these obstacles and develop a high switching frequency, high efficiency solution to achieve high power density and low profile, following techniques need to be improved: Advanced devices and material: at high switching frequency, the size of the magnetic component is limited by the magnetic loss. With better magnetic material, the size of the magnetic components could be significantly reduced. Loss on power semiconductor is the biggest part in total system loss. With better devices like CoolMOS, this loss could be reduced so that less thermal stress is imposed to the thermal management. Advanced packaging techniques: this includes advanced passive packaging and advanced active packaging. With advanced packaging technique, several

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aspects of improved could be expected. First, the utilization of space could be improved. With integration of capacitor into magnetic components, the total volume is greatly reduced. Second, with advanced packaging of active devices, the electrical performance of the system could be improved which will results to higher efficiency, lower noise. Third, advanced packaging could provide better thermal performance, which will help to reduce the volume of thermal management. Advanced power converter topology: for those state of the art topologies, high turn off loss and low efficiency at normal operating condition limits the ability of higher switching frequency and efficiency. With more advanced topology, switching loss need to be minimized so that high switching frequency could be achieved with high efficiency. Also, a desired topology need to be able to be optimized at high input voltage so that hold up time requirement will not impose serious penalty.

1.3 Outline of dissertation This dissertation is divided into five chapters. They are organized as following. First chapter is background review of 1kW front-end DC/DC power converter in Distributed Power System. The trends for this application are high power density, high efficiency and low profile. To achieve these targets, high switching frequency, high efficiency topology needs to be developed. Soft switching

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topologies like phase shift full bridge and asymmetrical half bridge has been the standard industry practice. For these topologies, the switching frequency is pushed to 200kHz. Then the turn off loss will be so high that increase frequency will not improve the power density. To achieve the future target, higher switching frequency is a must. This calls for more advanced technology. The primary target of this dissertation is to develop technique to achieve high frequency, high efficiency front-end converter that could be optimized at high input voltage while be able to cover wide input range. Chapter 2 presents several techniques developed to improve the performance of state of the art topologies. One problem for the state of the art topologies is the hold up requirement. With hold up requirement, front-end DC/DC converter needs to be designed with wide input range. Within the range, only the performance at high input voltage is critical. Unfortunately, for all those topologies, wide range design always penalizes the performance at high input voltage. Range winding solution and asymmetrical winding asymmetrical half bridge are two methods developed for wide input range issue. Quasi Square Wave synchronous rectifier is developed to reduce the secondary side conduction loss. Range winding solution provides the best performance possible for the state of the art topologies. Adding extra winding and devices could divide wide input

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range divided into two ranges. The converter could be optimized for a narrow range and use range winding to deal with wide input range. This method can be applied to any topology. Although range winding solution is effective, it needs extra windings, diodes, switches and control circuit. For asymmetrical half bridge converter, asymmetrical winding asymmetrical half bridge is a simpler and effective solution. With this solution, the duty cycle at high input voltage could be extended. With extended duty cycle, the voltage stress of output rectifier diodes could be reduced. With lower voltage rated diodes, the conduction loss and switching loss could be reduced greatly. This method is simple and effective, but it cannot be extended to other topologies like phase shift full bridge. Also, the output current will be discontinuous with this method. Quasi-square-wave synchronous rectification is a method to implement synchronous rectifier in front-end application. Compare 200V diode and MOSFET, 50% reduction of conduction loss could be achieved at 20A output current. For 300V MOSFET, the benefit will be very limited. To use 200V devices, symmetrical half bridge is chosen. Two issues make the result not so promising: body diode reverse recovery problem of synchronous rectifier and hard switching of primary switches. Quasi-square-wave operation mode solved these two problems with minimized penalty. With QSW operation, the body diode of synchronous rectifier will never conduct, which totally prevented the reverse

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recovery problem. Secondary inductor current also helps the primary switches to achieve zero voltage switching at whole load range. Although these techniques could improve the performance of the state of the art topologies, the high stress on the power devices and high switching loss problem is not been answered yet. Chapter 3 investigated advanced packaging technology, which could help reduce the stress and loss due to parasitic components. Passive integrated technology is also been discussed which could provide significant improvement on power density of passive components. Two integrated power electronics modules were developed for front end DC/DC converter: active IPEM, which integrated totem pole switches and drivers; passive IPEM, which integrated all the passive components in front end DC/DC converter except the output filter capacitor. With IPEMs, the power density of front end DC/DC converter is improved by more than 3 times. With advanced packaging technology, the performance of front end DC/DC converter could be improved, yet still the high switching loss and hold up time problem impose huge penalty on front end DC/DC converter design. Advanced topology still is needed to solve these two problems. Chapter 4 investigated the resonant topologies for this application. First many traditional resonant topologies are been investigated. They are Series Resonant

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Converter (SRC), Parallel Resonant Converter (PRC) and Series Parallel Resonant Converter (SPRC). All three topologies possess same problem as those PWM topologies: performance cannot be optimized at high input voltage for wide input range. At high input voltage, circulating current and switch turn off current reaches maximum. Another resonant topology: LLC resonant converter, fortunately, seems to be exactly what fits this application. With LLC resonant converter, first the circulating energy is minimized at high input voltage. The turn off current of switches is controllable and could be minimized. This topology is also capable of cover wide input range. Compare with state of the art topologies, 3% improvement of efficiency could be achieved. It is almost 40% reduction of the loss. With DC analysis, the operating region and design of LLC resonant converter is presented. Chapter 5 discussed two improvements for LLC resonant converter. To meet the profile and power density challenges, magnetic design is the most critical part in the system. A novel integrated magnetic structure is presented for LLC resonant converter. With integrated magnetic, all the magnetic components of a LLC resonant converter are integrated into one magnetic core. High power density can be achieved. Overload protection is another problem been addressed in this dissertation. To make practical use of this topology, methods to deal with abnormal situation is as 15

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important as the efficiency. Three methods to deal with over load situation were discussed. With increase switching frequency, the output current could be limited with the penalty of larger magnetic core size. With hybrid control of PWM and variable frequency control, previous problem could be prevented. The problem is lost of soft switching capability. The last method, which is a modified LLC resonant topology by adding clamp diodes to the resonant capacitor, can effectively control the output current with fast response. With above analysis, an open loop LLC resonant converter could be designed work very well. Next step is to close the voltage feedback loop. To do this, an understanding of the small signal characteristic of LLC resonant converter is essential. Chapter 6 is dedicated to the small signal modeling of LLC resonant converter. Traditional state space averaging method can no longer apply for resonant converter. Several different methods have been reported for this topic. Many of them made lot of simplifications, which makes the result not accurate. In this dissertation, two methods are used to extract the small signal model of LLC resonant converter. One method is based on simulation. It treats the converter as a black box. By inject small signal perturbation and monitor the output response at different perturbation frequency and operating point, a complete small signal characteristic of LLC resonant converter could be gained. This method is easy to implement and accurate as long as the circuit model is accurate. The drawback of

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this method is lack of intuition and time consuming. Another method is used as a complementary to the simulation method: extended describing function method proposed by Dr. Eric Yang. It is a general modeling tool for periodical operating system based on COSMR software package. With this method, a small signal model could be derived with zeros and poles identified. Compare these two methods, ETD method is fast but need state space model of the circuit in every situation while simulation method is accurate and easy but time consuming. Finally, based on the information gathered from above analysis, the feedback control could be designed for LLC resonant converter. With these analysis and test verification, LLC resonant converter is been proved to be an excellent candidate for front-end DC/DC conversion. The analysis and design are also been explored, although this exploration is far from completion, it enables industry to appreciate this topology as a possible for next generation front-end DC/DC converter.

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Chapter 2. State of the Art Topologies and Improvements

Chapter 2. State of the art topologies and improvements

2.1 Introduction In the first chapter, the trends for distributed power system were discussed. High power density, high efficiency and low profile are the key driving forces for technology development for this application. To achieve these targets, high switching frequency and high efficiency power conversion is necessary. In this chapter, first three state of the art topologies will be evaluated in detail. The issues of these converters will be discussed for this application. For the state of the art topologies, it almost reached the limit along this path with current technology. Switching loss and wide input range put lot of burden on these topologies, which prevented these topologies from increasing switching frequency and reaching higher efficiency. Several techniques will be developed to improve the state of the art topologies. First two techniques, range winding technique and asymmetrical winding asymmetrical half bridge technique, are developed to deal with wide input range problem caused by hold up requirement.

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For this application, with output current around 25A, secondary rectifier conduction loss is the biggest part in the total loss. Quasi Square Wave (QSW) Synchronous Rectifier technique is developed to reduce the secondary rectifier conduction loss. Range winding solution is a universal solution for wide input range problem. The concept could be implemented to any isolated topologies. For asymmetrical winding solution, it is limited to asymmetrical half bridge converter only. With modified control scheme, it could also be used for full bridge converter. For QSW synchronous rectification, currently it is demonstrate with symmetrical half bridge, it could also be used for phase shift full bridge, asymmetrical winding asymmetrical half bridge and range winding solution. With the fast advance in power MOSFET technology, it could also be extended to asymmetrical half bridge if 300V MOSFET could outperform 300V diodes.

2.2 State of the art topologies and issues 2.2.1 Two-switch-forward converter The schematic and waveform of two-switch forward (2SF) converter is shown in Figure 2.1 [A9][A10][A11]. The operation of two-switch-forward converter could be divided into three modes: energy transfer stage, transformer reset stage and dead time stage. In energy transfer stage, both primary switches are turned on; energy is transferred from input to output. In transformer-reset stage, two primary diodes will conduct and apply reversed input voltage to the transformer

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winding to reset the transformer. When transformer is totally reset, converter will come into dead time stage with no current in the primary side while secondary side is freewheeling. Two-switch forward got lot of interests and appreciations because of one reason: robust. Since two primary switches are not connected in totem pole configuration, they are turned on at the same time. This solved shoot through problem. For Half Bridge and Full Bridge converter, the primary switches are connected in totem pole structure. Whenever the two switches are turned on at the same time due to electromagnetic noise or radiation, it will be a destructive failure. For two-switch forward, this problem is solved, which is very critical for airspace power supply since they will be exposed to high-energy radiation. But to get this benefit, high price need to be paid.

Figure 2.1. Two-switch forward converter and operating waveforms

The major disadvantages of two-switch forward are hard switching and large filter inductor.

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Low efficiency is due to several reasons. First, two-switch forward is a hard switching converter; the switches are hard turn-on and turn-off. This will increase the switching loss for high frequency operation. Second, two-switch forward has higher conduction loss compare with half bridge and full bridge converters. This is because the energy transfer only happens during two switches are on. Because of transformer reset requirement, the maximum duty cycle can only reach 0.5. Which means at best, only half of the time this converter can transfer energy to the output; this will increase the RMS current through the primary switch. With same reason, the voltage-second on output inductor is much higher in two-switchforward converter compared with half bridge and full bridge converter. Because of these penalties of two-switch forward, it is not so widely used for this application now. For some application requires high reliability applications, it still been considered.

2.2.2 Phase Shift Full Bridge converter The schematic and operating waveforms of phase shift full bridge converter are shown in Figure 2.2 [A12][A13][A14][A15][A16]. Phase shift full bridge converter, as one of the most popular topology for this application, has many good characteristics. It is a soft switching converter. All four switches on primary side can achieve Zero Voltage Switching (ZVS) with proper design. This is very helpful for high frequency operation. This topology has lower volt-sec on the output filter inductor. Phase shift full bridge can achieve smallest volt-sec for

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same design specification compared with two-switch forward and half bridge converter.

Figure 2.2. Full bridge converter and operating waveforms

Another benefit of phase shift full bridge is its capability to cover wide power range. For power from several hundreds watts to kilowatts, full bridge converter can perform very well. In recent years, even for low power application like Voltage Regulator Module, full bridge topology is been investigated and showed benefits. There are several disadvantages for phase shift full bridge though. First, it is more complex than the other two topologies. With four switches on primary side, control and driver circuit will be more complex. Another problem is the leakage inductance. To achieve ZVS, large leakage inductance is needed. With large leakage inductance, the duty cycle loss due to charge and discharge leakage inductance will be significant. This will limit the choice of transformer turns ratio, which will affect the performance of whole converter. Even with large leakage inductance, still ZVS cannot be achieved at light load. There are many papers 22

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discussed different methods to expend ZVS region for phase shift full bridge, but they are not been widely adopted due to complexity. Another issue for phase shift full bridge is the circulating current. As seen in the waveforms, during every switching cycle, there is a time interval during which two up switches or low switches are turned on at the same time. This will short the transformer primary side. During this time interval, secondary is freewheeling and no energy transfer from input to output. Primary current during this period is pretty high. This current circulates through the primary two switches and transformer winding. It will increase the conduction loss. Smaller the duty cycle, more circulating current will be. Although some drawbacks for phase shift full bridge, it is still a popular topology for this application. Its capability to operate at high frequency and wide power range enable it to be used for multi applications.

2.2.3 Half bridge converter As shown in Figure 2.3, half bridge converter has only two switches on the primary side [A17][A18][A19][A20][A21]. For half bridge converter, with different control signal, the converter operates very differently. The different operating waveforms are shown in Figure 2.4 and Figure 2.5. When the two switches are driven with symmetrical signals, which are identical to each other with 180-degree phase shift, the converter is called symmetrical half bridge. For the one with complementary driving signals, it is called asymmetrical half bridge. 23

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Figure 2.3. Schematic of half bridge converter

For symmetrical half bridge, it is hard switching topology. Leakage inductance is detrimental to the performance of converter. Normally snubber circuit is needed to absorb the ringing problem caused by leakage inductance during the period when both switches are off. For symmetrical half bridge, the power level it limited because of these limitations.

Figure 2.4. Operating waveforms of symmetrical half bridge converter

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Figure 2.5. Operating waveforms of asymmetrical half bridge converter

Asymmetrical half bridge is a very interesting topology. It has some unique characteristics. First, asymmetrical half bridge is a soft switching converter. Primary two switches can achieve ZVS with help of leakage inductance. Since the two switches works complementarily, there is no ringing problem caused by leakage inductance. Also, since energy is transferred from input to output during whole switching period, there is no circulating current as seen in phase shift full bridge. For asymmetrical half bridge, there are several drawbacks too. One problem is that the voltage stress on the secondary rectifier is asymmetrical and related to duty cycle. In some situations, the voltage stress on the output filter diodes could reach very high, which will limit the choice of diodes. Since asymmetrical half bridge also utilizes leakage inductance to achieve soft switching, there is similar problem as discussed for phase shift full bridge, which is lost of ZVS during light load condition. 25

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For asymmetrical half bridge, the transformer is biased according to different duty cycle. So the design of asymmetrical half bridge transformer need take this into consideration. Asymmetrical half bridge is very popular for power level less than 1kW application because its simplicity compared with phase shift full bridge. Current doubler is a topology for secondary rectifier. It is widely used for low voltage, high current applications because of reduced transformer winding loss and easy to implement magnetic integration concept. For front-end DC/DC application, the secondary current can reach 25A. With current doubler and magnetic integration, higher efficiency and higher power density can be achieved.

Figure 2.6. Asymmetrical half bridge with current doubler and waveforms

For asymmetrical half bridge with current doubler, its operation is same as traditional asymmetrical half bridge. For current doubler asymmetrical half bridge, since the two inductor Lf1 and Lf2 current could be different, the transformer doesn't need to be biased. With current doubler structure, the transformer loss could be reduced. 26

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2.2.4 Comparison of three topologies Figure 2.7 shows the loss comparison of these topologies for primary switches. Two switch forward and symmetrical half bridge converters have much higher loss compared with full bridge and asymmetrical half bridge converter. Since two-switch forward and symmetrical half bridge converter are hard switching converters, the operating frequency cannot be pushed too high. This will limit the achievable power density.

Figure 2.7. Loss comparison of three state of the art topologies

Figure 2.8 shows the comparison of volt-second on output filter inductor for these topologies. For half bridge and full bridge, when duty cycle reaches maximum, the volt-sec on the output filter inductor will be very small, this will reduce the required inductor size if same current ripple is assumed. For two switch forward, however, even with maximum duty cycle, the volt-sec is still very large due to the discontinuous energy transfer property of the converter. As seen

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in Figure 2.8, two-switch forward converter has almost 5 times volt-sec on the output filter inductor compared with half bridge and full bridge converter.

Figure 2.8. Output filter inductor volt-sec comparison of three topologies

From above comparison, two-switch forward and symmetrical half bridge converter are hard switching converters, which are not suitable for high frequency operation. For full bridge converter, and asymmetrical half bridge converters, they both have pros and cons. Full bridge converter could provide some improvement on output filter inductor volt-sec, but high circulating current and complex structure are its drawbacks. Half bridge converter has simple structure, although the volt-second on output filter inductor is a little bit higher than full bridge. In next part, the impact of hold up time on these two converters will be discussed.

2.3 Issue with hold up time requirement Hold up time requirement is one special requirement for front-end application. It requires the front-end system to provide full power output for one AC line

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cycle, which is 20ms ,after the AC line is lost. This hold up time will ensure the digital system to have enough time to respond to power failure. With 1kW output power and 20ms hold up time requirement, the energy needed to store in the system will be at least 20Joule. Observing the system, there is no much choice but to use intermediate 400V DC bus capacitors as the energy storage component. This is because of several reasons. First, this intermediate bus is highest voltage bus; it is more efficient to put capacitor here to store more energy. Second, this bus voltage is loosely regulated; it can have a large variation. For 48V output, it needs to be tightly regulated, so it is not possible to use energy on the 48V DC bus capacitors. During hold up time, energy is drawn from the 400V DC bus capacitor. With energy drawn from it, the voltage on these capacitors will drop. After the hold up time, the voltage on the bus capacitor will be much less than normal operating voltage that is 400V. This will be a design trade off to be made. With smaller bus capacitor, this variation will be larger. Although the size of the bus cap can be reduced, the performance of the front-end DC/DC converter will be greatly penalized due to the wide input range. While with big bus capacitor, the performance of front-end DC/DC could be optimized; the size and cost of bus cap will be a penalty. In Figure 2.9, the relationship between capacitance needed for hold up time and lowest input voltage the bus will drop after hold up time is showed. For 400V 29

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DC bus, 450V electrolytic capacitor is used. For this voltage rated capacitor, 330uF is most cost and volume effective. The choice will be how many 330uF capacitors will be needed.

Figure 2.9. Capacitance for hold up time and DC bus voltage after hold up time

Table 2-1 shows the capacitance used and the input voltage range for the front-end DC/DC converter. It is easy to see that 660uF is a good choice, which is also the value used by many industry products. Table 2-1. Input range of front-end DC/DC converter vs. DC bus capacitance

With 660uF bus capacitor used, the input voltage range of the front-end DC/DC converter will be 300 to 400V. During the whole lifetime of the converter, it worked with input voltage around 400V. Only in very rare power

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failure, the converter will work down to 300V input. The question is how much penalty is paid for 400V input operation in order to operate the converter at 300V input. Next part asymmetrical half bridge and phase shift full bridge will be analyzed on this aspect.

2.3.1

Asymmetrical half bridge with hold up time design

For asymmetrical half bridge, the maximum duty cycle is 0.5. When duty cycle is 0.5, the converter reaches highest gain too. Because of the duty loss due to leakage inductance, the maximum achievable duty will be set to 0.45.

Figure 2.10. Duty cycle range for asymmetrical half bridge with hold up requirement

To cover input voltage from 300V to 400V, the highest gain need to be designed at 300V. With duty cycle equals to 0.45, transformer turns ratio is decided accordingly. When input voltage increase to 400V, duty cycle will reduce to reduce the gain and keep output voltage regulated. The duty cycle will be less than 0.25 when input voltage is 400V as shown in Figure 2.10. From this we can

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see, to design with hold up time requirement, the duty cycle of the converter will be very small during normal operation. Next, the impact of small duty cycle at normal operation on the performance will be discussed. For asymmetrical half bridge shown in Figure 2.6, the secondary side rectifier diodes voltage stress will be related to input voltage and duty cycle as:

VD1 =

ns ns ⋅ (1 − D) ⋅ Vin and VD 2 = ⋅ D ⋅ Vin . np np

Average current flow through each diode will be: I D1 = (1 − D) ⋅ I O and I D 2 = D ⋅ I O

Figure 2.11. Secondary rectifier voltage and current stress for asymmetrical half bridge

As seen from Figure 2.11, when input voltage is 400V, the voltage stress for D1 reaches maximum because of high input voltage and small duty cycle. Also, most of the load current will be carried by D1 too. The maximum voltage is over

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210V, so 300V diodes has to be used for the secondary rectifier. The highest voltage rating shottky diode available is 200V. With 300V diodes, the forward voltage drop and reverse recovery characteristic is much worse than 200V shottky diode. This will increase the conduction and switching loss.

Figure 2.12. Test efficiency of asymmetrical half bridge

Figure 2.12 shows the test efficiency of asymmetrical half bridge for two cases. In first case, the converter is designed just to operate from 360V to 400V. In the other case, the converter is designed according to hold up time requirement with 300V to 400V input range. The differences between these two designs are shown in Table 2-2. Table 2-2 Difference between two asymmetrical half bridge design Input Range

Xfmer turns ratio

Case 1

360-400V

10:6

Case 2

300-400V

10:7

Secondary diode

Primary Switch

MUR2002 200V Shottky Diode STTH3003 300V Fast recovery diode

IXFH22N50 500V 20A IXFH22N50 500V 20A

33

Switching Frequency 200kHz 200kHz

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From test results we can see, because of this wide input range, efficiency at 400V drops more than 3%, which means more than 60% increase in loss because of wide input voltage range.

2.3.2

Phase shift full bridge with hold up time design

For phase shift full bridge, with hold up time requirement, its duty cycle range is shown in Figure 2.13. When input voltage drops to 300V, duty cycle reaches the maximum. When input voltage is 400V, duty cycle is only around 0.34.

Figure 2.13. Duty cycle range for full bridge converter with hold up requirement

For phase shift full bridge, the secondary rectifier voltage stress is not a big concern. The major problem is the high circulating current during normal operation. As seen in Figure 2.14, during each switching cycle, there is a free wheeling period. Secondary current freewheels through secondary diodes in this time interval. Primary also has high current circulating through primary two switches. This primary circulating current will increase conduction loss and

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should be minimized to achieve high efficiency. Unfortunately, to design for hold up time, the duty cycle at 400V is very small, less than 0.35. This means during each switching cycle, the current circulating for almost one third of the time.

Figure 2.14. Circulating current of full bridge with different input voltage

2.3.3

Conclusion

From above analysis, following conclusion could be drawn: 1. For front-end DC/DC application, to design for hold up time, its input voltage range will be wide. With 660uF bus capacitor, the input voltage of frontend DC/DC converter will be 300 to 400V. 2. For phase shift full bridge, to design for this specification, more than one third of the primary current is circulating without transfer energy. 3. For asymmetrical half bridge, the secondary rectifier voltage stress will increase to very high due to wide input range requirement. 4. From test results, it can be seen that more than 60% loss increase will occur because of wide input range requirement. 35

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A desired front-end DC/DC converter should have following characteristic: • Could be optimized at high input voltage while still cover wide input range • High efficiency and high switching frequency, so that high power density and low profile could be achieved with high switching frequency.

2.4 Range winding for wide input range As discussed in the previous part, to deal with hold up time requirement, front end DC/DC converter need to cover wide input range. Within the input range, the converter will work at high input voltage for the whole lifetime. Only during 20ms hold up time, the converter will operate at low input voltage down to 300V. The performance at high input voltage is critical to system while the performance at hold up time is not a consideration as long as the converter still functional. Unfortunately, the PWM converter is working at better condition with low input voltage. As input voltage increases, duty cycle will reduce and the converter is working in worse condition. To design for wide input range, the efficiency at high input voltage will drop significantly. As shown in first chapter, with wide input range design, the efficiency at high input voltage is around 92%. If the converter is designed just for a narrow input range, the efficiency can reach 95%. Range winding solution is a method to deal with this problem. The concept of range winding solution is to change the transformer turns ratio according to different input voltage so that the transformer could be optimized for high input

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voltage. In the following part, asymmetrical half bridge with range winding will be discussed as an example to demonstrate the operation of range winding concept.

Figure 2.15. Asymmetrical half bridge converter with range winding

As shown in Figure 2.15, range-winding solution is built upon original asymmetrical half bridge converter by adding extra windings, diodes and switch. By adding those extra components, another freedom is added to the converter. With turn on or off the range switch Qr, the transformer winding turns ratio could be changed. In this way, the gain of the converter could be regulated through two ways, duty cycle and range switch. When range switch Qr is turned off, the converter will have a gain:

Vo =

Vin ⋅ (1 − D) ⋅ D ⋅ nS , Which is same as an asymmetrical half bridge with nP

transformer turns ratio np: ns. With the range switch turned on, the range winding will be added to the secondary, the converter will have a different gain: 37

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Chapter 2. State of the Art Topologies and Improvements

Vin ⋅ (1 − D) ⋅ D ⋅ (n S + nsr ) nP

Figure 2.16 Circuit diagram for normal operation

Figure 2.17 Circuit diagram for hold up operation

As shown in those equations, with range winding turned on, the transformer turns ratio will be reduced. The converter will have higher gain with lower turns ratio. By detect the input voltage of front end DC/DC converter, when it drops below given level, the range switch will be turned on, and the converter will have higher gain to cover lower input voltage. During normal operation, the range switch is turned off, the range winding and diodes will not affect the operation.

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With range winding solution, the inner transformer turns ratio (np:ns) could be optimized for high input voltage so that the converter will operate with large duty cycle at normal operation. At hold up time, the range switch will be turned on to increase the gain to cover wider input range. To demonstrate the benefits of this solution, design examples of traditional asymmetrical half bridge and asymmetrical half bridge with range winding are performed and compared through analysis and experiments. For a traditional asymmetrical half bridge, to design for input voltage range from 300V to 400V, the transformer turns ratio is choose to be 10:7 (np:ns). With this turns ration, the duty cycle at 300V will be 0.45. The maximum duty cycle is set to 0.45 considering the duty loss caused by leakage inductance for soft switching. When input voltage is 400V, duty cycle is less than 0.25. For asymmetrical half bridge with range winding, the transformer turns ratio (np:ns) is designed to be 10:6. The range winding nsr is just one turn. With turn ratio 10:6, the converter will be able to cover a input range from 360V to 400V. At 400V, the duty cycle is 0.34. When input voltage drops to below 360V, range switch will turn on and the converter will have same gain characteristic as traditional asymmetrical half bridge with turns ratio of 10:7. Next the effect of range winding will be explored.

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Figure 2.18 Duty cycle range comparison of asymmetrical half bridge with/without range switch

In Figure 2.18, the duty cycle of two converters at different input voltage are compared. For traditional asymmetrical half bridge, the duty cycle at 400V is less than 0.25. With range winding, the duty cycle is extended to 0.34, which is about 40% improvement over traditional asymmetrical half bridge. With extended duty cycle, the stress on all components will be changed. First, as shown in Figure 2.19, the voltage stress on the rectifier diodes is reduced with range winding solution. Without range winding, the voltage stress on the rectifier diodes is more than 210V, normally 300V diode has to be used. For range winding solution, the voltage stress on the range winding is reduced to 160V; 200V shottky diode can be used. With shottky diode, forward voltage drop and reverse recovery loss are reduced significantly. For traditional asymmetrical half bridge, the secondary conduction loss is around 25W at full load condition. For range winding asymmetrical half bridge, the secondary conduction loss is reduced to 17W with shottky diodes. In traditional asymmetrical half bridge, the diode reverse recovery problem is very severe, snubber circuits have to be added 40

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to reduce the ringing caused by reverse recovery current. With shottky diodes, the reverse recovery problem could be eliminated and snubber could be eliminated. This will improved the efficiency at high switching frequency. As shown in Figure 2.20, the current stress on the two rectifier diodes are also more balanced during normal operation condition, which is beneficial for thermal design.

Figure 2.19 Diode voltage stress comparison

Figure 2.20 Diode average current comparison

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Another benefit comes from the decreasing of primary RMS current and turn off current with extended duty cycle. As shown in Figure 2.21 and Figure 2.22, the RMS current and turn off current of the primary switches are reduced with range switching solution. With reduced RMS current, primary conduction loss could be reduced. With reduced turn off current, switching loss could also be reduced.

Figure 2.21 Primary switch RMS current comparison

Figure 2.22 Primary switch turn off current comparison 42

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Test circuits were built for these two cases. In Figure 2.24, the test efficiency for both converters is shown at normal operation condition.

Figure 2.23 Test setup for range winding asymmetrical half bridge

Figure 2.24 Test efficiency at normal operation for range winding solution

From the test efficiency result, it can be seen that range winding can improve the efficiency at normal operation condition significantly. The drawback is extra winding on the transformer and several extra devices.

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Range winding concept could also be extended to other topologies like full bridge, and current doubler structure as shown in Figure 2.25.

(a)

(b)

Figure 2.25 Range winding for (a) Full bridge (b) current doubler

2.5 Asymmetrical Winding Asymmetrical Half Bridge With range winding solution, the performance of front end DC/DC converter could be improved significantly at high input voltage condition. This method could be extended to other isolated DC/DC topologies. The drawback of range

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winding solution is that extra windings, diodes, switch and control circuit are needed, which makes this solution very complex. For asymmetrical half bridge with current doubler, another simpler yet effective solution is possible: asymmetrical winding asymmetrical half bridge. For asymmetrical half bridge with current doubler, it has one transformer and two inductors as shown in Figure 2.26. This topology could be transformed into a two-transformer version with same operation and characteristic. The transform is demonstrated in Figure 2.27. Each step of the transformation is reversible, so these two topologies are exactly the same when the two transformers have same turns ratio as the single transformer.

(a)

(b)

Figure 2.26 Half bridge current doubler with (a) one transformer and (b) Two transformers

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Figure 2.27 Topology transformation of current doubler

With two transformers, another dimension of freedom is provided to the designer. With different turns ratio for these two transformers, some interesting phenomenon could be observed. For traditional asymmetrical half bridge, the equation for duty cycle in relationship with input, output voltage and turns ratio is:

Vo =

Vin ⋅ (1 − D) ⋅ D np

For two-transformer asymmetrical half bridge, the duty cycle will be:

Vo =

Vin ⋅ (1 − D) ⋅ D n p1 ⋅ D + n p 2 ⋅ (1 − D)

These equations are drawn in Figure 2.28. When the two transformers have same turns ratio, maximum gain is achieved at duty cycle equals to 0.5. This is same characteristic for one transformer asymmetrical half bridge. When the two transformers have different turns ratio, the maximum gain will be shifted from duty cycle equals to 0.5. When np1 is larger than np2, the maximum gain will

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shift to smaller duty cycle. When np1 is smaller np2, the maximum gain will shift to larger duty cycle.

Figure 2.28 DC characteristic for different np ratio

The design goal for front end DC/DC converter is to optimize the performance at high input voltage. To achieve this goal, we would like to extend the duty cycle at high input voltage. From this prospective, np1 smaller than np2 is preferred. With np1 smaller than np2, the maximum gain will shift to duty cycle larger than 0.5 and the DC characteristic will tilt toward right. With this effect, the duty cycle at high input voltage will be extended too. The duty cycle for the single transformer asymmetrical half bridge is:

D=

Vin − Vin2 − 4 ⋅ Vin ⋅ Vo ⋅ np 2 ⋅ Vin

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The duty cycle for two transformers asymmetrical half bridge is:

D=

Vin + Vo ⋅ (np 2 − np1) − Vin2 + Vo2 ⋅ (np 2 − np1) 2 − 2 ⋅ Vin ⋅ Vo ⋅ (np1 + np 2) 2 ⋅ Vin

In Figure 2.29 duty cycle range for different np1 to np2 ratio are shown. With smaller np1 to np2 ratio, the duty cycle at 400V could be shifted closer to 0.5. Now the question is what ratio of np1 to np2 should be chosen. To answer this question, the impacts of different np1 to np2 ratio need to be investigated. In the following part, the secondary diode voltage stress, current stress, output current ripple and inductor current will be analyzed.

Figure 2.29 Duty cycle range with different turn ratios

For asymmetrical half bridge, because of the high voltage stress on the rectifier diode, high voltage rating diodes have to be used. This increases the conduction loss and switching loss of secondary rectifier significantly. In fact, the 48

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secondary rectifier contributes almost half of the total loss. If the voltage stress of secondary diodes could be reduced so that lower voltage diodes could be used, the efficiency could be improved significantly. For two transformer asymmetrical winding asymmetrical half bridge, the voltage stresses of the secondary diodes are:

VD1 =

D ⋅ Vin np 2 + Vo ⋅ (1 − ) np1 np1

VD1 =

(1 − D) ⋅Vin np1 + Vo ⋅ (1 − ) np 2 np 2

In these equations, when np1 equals to np2 is used, it will be the voltage stress for traditional asymmetrical half bridge. Next the current of the two current doubler inductors will be analyzed. From the charge balance for the primary DC blocking capacitor, the current of two output inductors could be derived as:

I L1 =

D ⋅ I o ⋅ np1 D ⋅ np1 + (1 − D) ⋅ np 2

I L2 =

(1 − D) ⋅ I o ⋅ np 2 D ⋅ np1 + (1 − D) ⋅ np 2

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(a)

(b)

Figure 2.30 Operation waveforms of (a) asymmetrical half bridge and (b) asymmetrical winding asymmetrical half bridge

I1 = I L 2 +

np1 I L1 np 2

I 2 = I L1 +

np 2 I L2 . np1

Above I1 and I2 is also the current stress on the two output diodes. It can be seen that the two diodes have very different current stress. One diode will see very high current while the other one only pass through very small current. Above equations are drawn in Figure 2.31 to Figure 2.33. 50

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Figure 2.31 Diode voltage stress for asymmetrical winding asymmetrical half bridge

Figure 2.32 Diode current stress for asymmetrical winding asymmetrical half bridge

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Figure 2.33 Output current ripple for asymmetrical winding asymmetrical half bridge

From above analysis, we can see that the voltage stress on the output diodes is reduced with asymmetrical winding solution. The penalty is the discontinuous output current and unbalanced current stress on the output diodes. This unbalance will also show in the power transfer ratio of the two transformers. By balancing all these aspects, np1 to np2 ratio of 1:3 is chosen. With this ratio, the voltage stress of two diodes could be limited below 150V so that 200V shottky diodes could be used. Figure 2.34 to Figure 2.36 shows the test circuit, test waveform and efficiency. From the test result, efficiency could be improved by 1.5% with this solution. The performance improvement of this solution is not as good as range winding solution, but because of its simplicity, it is a good solution for asymmetrical half bridge converter.

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Figure 2.34 Test circuit diagram of asymmetrical winding asymmetrical half bridge

Figure 2.35 Test waveforms of asymmetrical winding asymmetrical half bridge

0.96

Traditional Asy. HB Asymmetrical Winding

Efficiency

0.95 0.94 0.93 0.92 0.91 0.9 0

5

10

15

20

25

Load Current (A) Figure 2.36 Test efficiency of asymmetrical winding asymmetrical half bridge 53

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2.6 QSW Synchronous Rectification 2.6.1 Synchronous Rectifier for front end DC/DC Converter

In low voltage, high current application, secondary rectifier conduction loss is dominant in total system loss. With advanced power MOSFET technology, the conduction loss of a MOSFET is much less than diodes. Synchronous rectifier has been a must technology for low voltage, high current application. For front end DC/DC application, output current is higher than 20A. With diode rectifier, the conduction loss is a big part in total converter loss too. For example, with 300V diode in asymmetrical half bridge, the conduction loss of secondary diodes is about 25W, which is more than 30% of total loss. In this part, the synchronous rectification for this application will be discussed. For front end DC/DC application, the voltage stress on the rectifier diodes is much higher than low voltage application. To verify the viability of synchronous rectification in front end DC/DC application, conduction loss for different devices will be compared first. For front end DC/DC converter, output voltage is 48V. The voltage stress for secondary diodes is shown in Figure 2.37. For asymmetrical half bridge, voltage stress on the diodes is higher than symmetrical half bridge and full bridge. With a safe margin, the device chosen for asymmetrical half bridge are 300V devices while for range winding, asymmetrical winding, symmetrical half bridge and full bridge, 200V devices could be used. Searching the available devices, following devices are been used for comparison:

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300V diode: STTH3003 from IXYS, Vf=1.2V with trr=40ns. 300V MOSFET: MTW32N25E from On-Semi, Rdson=80mohm. 200V diode: MUR202000, shottky diode with Vf=0.9V. 200V MOSFET: MTY55N20E from On-Semi,, Rdson=28mohm.

Figure 2.37 Rectifier diode voltage stress of different topologies

Figure 2.38 Conduction loss comparison of different devices

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Figure 2.38 shows the conduction loss for different devices with load change. As seen in the graph, 200V MOSFET could provide much improved performance with current in 20A range over 200V diode. For 300V devices, single MOSFET cannot provide much different, only with more MOSFET in parallel so that the conduction loss can be reduced. It doesn't make much sense to parallel two or more MOSFET here since the thermal is manageable with only one device. To get improved performance with same amount of devices, 200V will be more meaningful. With voltage stress shown in Figure 2.37, it can be seen that asymmetrical half bridge will not get much benefit from synchronous rectification. For asymmetrical winding asymmetrical half bridge, range winding, full bridge and symmetrical half bridge converter, 200V devices could be used. In these topologies, synchronous rectifier might be beneficial. For range winding solution, with synchronous rectifier, the converter will become very complex. Also the improvement will be limited since performance of 200V diode is not so bad. For asymmetrical winding asymmetrical half bridge, it is possible to use synchronous rectifier. The problem is the unbalanced current stress on the two rectifier diodes. As shown in Figure 2.39, two rectifier diodes see very different current stress. Without parallel devices, synchronous rectifier will not show much improvement over one diode. With two 200V MOSFET parallel as one

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synchronous rectifier, it is possible to reduce the conduction loss with the price of more devices as shown in Figure 2.40.

Figure 2.39 Rectifier diodes current stress for asymmetrical winding asymmetrical half bridge

Figure 2.40 Conduction loss comparison of 200V diode and MOSFET

For symmetrical half bridge and full bridge converter, they will be good candidates to demonstrate the benefits of synchronous rectifier. In this part, symmetrical half bridge will be used to demonstrate this technique as shown in

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Figure 2.41. All the method discussed here could be used for full bridge converter.

Figure 2.41 Circuit diagram of half bridge current doubler with synchronous rectifier

Figure 2.42 Test waveform of synchronous rectifier for front end converter'

Figure 2.42 shows the test waveform for symmetrical half bridge with synchronous rectifier. From the waveform, a huge current spike and voltage ringing could be observed. All these spikes are caused by the reverse recovery of body diodes of synchronous rectifier as shown in Figure 2.43

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The two synchronous rectifiers are working complementary with primary switches. Before Q2 is turned on, S1 has to be turned off first; otherwise a short circuit will exist. To guarantee the safe operation, a dead time is inserted between turning on of Q2 at t3 and turning off of S1 at t2. During the dead time, synchronous rectifier S1 is turned off. The current will flow through the body diode of synchronous rectifier S1. When Q2 is turned on at t3, the body diode of S1 will be forced turn off. Since the body diode of MOSFET is very slow, a huge reverse recovery current will appear. This reverse recovery current will also cause voltage ringing on the synchronous rectifier.

Figure 2.43 Operation waveform of synchronous rectifier for front-end converter

With the problem identified, the solution will be a method, which could prevent the conduction of the body diode of synchronous rectifier. Quasi Square

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Wave (QSW) synchronous rectification is proposed to solve this problem. The concept of QSW synchronous rectification is shown in Figure 2.44.

2.6.2

QSW Synchronous Rectifier

The idea is by reducing the filter inductor, the ripple current of each inductor will increase, at some point, the ripple will be large enough so that the direction of the current will change during each switching cycle. As seen in the graph, if the inductor current goes to negative, at time t2 when the synchronous rectifier S1 is turned off, the current is flowing through drain to source. This current cannot go through the body diode. So when synchronous rectifier is turned off, the body diode will not conduct. This way, the conduction of body diode is prevented.

Figure 2.44 Operation waveform of QSW synchronous rectifier

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Figure 2.45 Operating mode 1 of half bridge with synchronous rectifier

Figure 2.46 Operating mode 2 of half bridge with synchronous rectifier

Figure 2.47 Operating mode 3 of half bridge with synchronous rectifier

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Figure 2.48 Operating mode 4 of half bridge with synchronous rectifier

Figure 2.49 Operating mode 5 of half bridge with synchronous rectifier

Another benefit is Zero Voltage Switching. To use 200V MOSFET, symmetrical half bridge is used. With symmetrical half bridge, ZVS cannot be achieved. With QSW synchronous rectification, we can achieve ZVS for the primary switches. At time t2, synchronous rectifier S1 is turned off. Since the current is negative, it cannot go through the body diode. Then this current will be flow through transformer. This current then reflects to primary side of the transformer and it will discharge the output capacitor of primary switch Q1 and Q2 so that ZVS could be achieved.

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With QSW synchronous rectification, reverse recovery problem of body diodes of synchronous rectifier could be solved. Primary switches also can achieve ZVS. But to achieve QSW operation mode, the inductor current ripple will be very large. The question is how much increasing of conduction loss will be caused by large current ripple. For secondary two synchronous rectifiers, the current flowing through them is the sum of two inductor currents as seen in Figure 2.44. Since the two inductor currents are out of phase to each other. When they add up, the ripple will be much smaller. So for secondary side, the large current ripple will not show up. The question is the primary side two switches. For this technique, the primary switches will see the current ripple of each individual inductor. So conduction loss of primary switches will increase. But in this application, the primary conduction loss is not significant in total loss. As shown in Figure 2.50 although the primary conduction loss will increase about 35%, still better efficiency could be expected because of reduced conduction loss of secondary side and zero voltage switching for primary switches.

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Figure 2.50 Primary loss comparison of QSW and conventional synchronous rectifier

Figure 2.51 shows the test circuit of half bridge with QSW synchronous rectifier. It is working as symmetrical half bridge.

Figure 2.51 Test circuit of QSW synchronous rectifier

The test waveform is shown in Figure 2.52 for light load and in Figure 2.53 for heavy load. QSW operation mode could be identified from the negative current. There is no current spike in QSW operation mode. The efficiency test results are shown in Figure 2.54. With QSW synchronous rectification, the

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efficiency at full load could reach 95%. For synchronous rectifier without QSW, after push output current over 10A, the devices will be destroyed by excessive reverse recovery loss.

Figure 2.52 Test waveform of QSW synchronous rectifier at light load

Figure 2.53 Test waveform of QSW synchronous rectifier at heavy load

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Figure 2.54 Test efficiency of half bridge with QSW synchronous rectifier

QSW synchronous rectification could effectively eliminate body diode reverse recovery problem of synchronous rectifier. With current doubler output filter, the current ripple on synchronous rectifier will not increase significantly compared with non-QSW mode. This method could be implemented to other topologies too. Figure 2.55 shows the diagram of Phase shift full bridge with QSW synchronous rectifier. In other application where synchronous rectifier is used, this method could also provide a simple solution to the body diode reverse recovery issue.

Figure 2.55 Full bridge with QSW synchronous rectifier

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2.7 Summary and issues In this chapter, several techniques are presented to improve the performance of half bridge converter. The target is to improve the efficiency at normal operation mode.

Figure 2.56 Efficiency comparison of different methods

Range winding solution can effectively improve the efficiency at high input voltage and use range winding to cover wide input range. It enables the design to be concentrated only on the performance of our interest and gives best performance possible. However, to implement range winding solution, extra components are needed which makes this solution very complex. Asymmetrical winding solution is a simpler yet effective solution for asymmetrical half bridge converter. This method doesn't need extra components. With asymmetrical winding, the duty cycle at high input voltage could be extended. With extended duty cycle, the voltage stress of output rectifier could be reduced so that shottky diode could be used. With shottky diode, conduction loss

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and switching loss could be reduced. There are some side effects of this method. With asymmetrical winding, the output current will be discontinuous. This requires more output capacitor. Also, the current stress is unbalanced for secondary devices. In the design, this should be taken into consideration. The QSW synchronous rectification technique, which enables the using of synchronous rectification in this application, could reduce the conduction loss and also help primary switches to achieve zero voltage switching. With QSW mode, the body diode of synchronous rectifier will never conduct. Primary switches can achieve zero voltage switching at whole load range. Currently, because of the limitation of device, only when the voltage stress is lower than 200V, synchronous rectifier shows benefit over diode rectifier. These limits the topologies could use this technology. With more advanced power MOSFET technology, this limitation could be raised to high voltage. It will enable wider range of application for QSW synchronous rectification. The other problem with QSW synchronous rectifier is the high turn off current for primary switches, which will introduce high switching loss. This makes QSW synchronous rectifier not suitable for high switching frequency. Compare the efficiency and complexity, range winding provides highest efficiency at normal operation with relatively simple circuit. Although these methods could improve the efficiency of front-end converter at normal operation, to achieve high power density and low profile, high switching frequency operation is also necessary. But for both the state of the art

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topologies, switching loss is a big obstacle for high frequency operation. Here the switching loss just take turn off loss of primary power MOSFET into consideration. From Figure 2.57, if switching frequency increases to 400kHz, switching loss will be much higher. This will require more space for thermal management of the power devices. This is a big limitation for PWM converter to achieve high power density.

Figure 2.57. Primary switch total loss for different switching frequency

Figure 2.58 Half bridge converter with snubber circuit

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Figure 2.59 Test waveform with and without snubber circuit

Another problem for the state of the art topologies is the reverse recovery problem of secondary rectifier diodes. As seen in Figure 2.59, with diode reverse recovery, the current stress and voltage stress is a serious problem. With saturable core snubber shown in Figure 2.58, the problem could be reduced as shown in Figure 2.59. The problem is for snubber circuit; thermal problem and high loss prevent them to operate with high switching frequency. With 200kHz switching frequency, the temperature of the saturable core is already 85 degree.

Figure 2.60 Prototype of 200kHz asymmetrical half bridge with discrete components

Figure 2.60 shows the prototype of a 200kHz asymmetrical half bridge converter. Magnetic and capacitor occupied huge space in the system. Non uniform profile also wasted lot of space. In fact, the active switches and the 70

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driver also take lot of space in the system. In next chapter, the advanced packaging technology will be discussed for this converter. As will be demonstrated in next chapter, with advanced packaging technology, the power density of the converter could be significantly improved.

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Chapter 3. Integrated Power Electronics Module

Chapter 3 Integrated Power Electronics Module 3.1 Introduction To achieve high power density, low profile, the fundamental approach of electrical power processing is steadily moving toward high switching frequency. From previous analysis, high switching loss and high stress limited the ability to operate front end DC/DC converter at higher switching frequency. From the history of power electronics, the technology improvements in power semiconductors have been the driving force for this problem. Moving from bipolar to MOSFET technology has greatly increased switching speed. For current semiconductor technology, very fast devices are available. For the power MOSFET used in front end application, they could be switched with mega hertz frequency range. The limiting factor now is the packaging technology. Because of the parasitic inductance and capacitance due to the packaging technology, the switching action has to be slowed down to limit the stress and undesirable noise problem. This limited the ability to reduce the switching loss of the devices. For a typical front-end converter, individual power devices are mounted on the heat sink. Driver, sensors and protection circuits are implemented on a PCB, which is mounted close to power devices. The power devices are packaged with wire bonding technology. This kind of package has several limitations: first, the 72

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parasitic related to the wire bond and PCB connection is very large; second, with wire bond technology, three-dimensional integration is not possible which limits the electromagnetic layout and thermal management. Some manufacturers have taken a more aggressive approach by integrate power semiconductors in die form on one common substrate with wire bonding. This approach provides some improvement by putting devices closer, but it doesn’t resolve the limitations of traditional packaging technology. Figure 3.1 shows the circuit diagram and switching waveform of a totem pole switches with interconnect parasitic considered with wirebond technology. When the MOSFET is switched at high current, those parasitic will introduce high stress on the devices. High voltage spike and ringing exists when turning off the switch. As switching speed increasing, the voltage overshoot will increase.

Figure 3.1 Totem pole switches with parasitic inductance and Q2 Vds (100V/div)

Another problem comes from the gate related parasitic. As shown in Figure 3.2, the gate driver loop shares Ls with power path. This will greatly impact the switching performance of switches. During turn off of upper switch Q1, Ls will 73

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introduce a voltage VLs in the gate loop. This voltage will slow down the turn off speed thus introduce higher switching loss.

Figure 3.2 Impact of gate loop parasitic

Other than these electrical performance impacts of current packaging technology, thermal and noise problem will also greatly impact the capability of these technology for higher switching frequency. From above discussion we can see, while power devices are still one of the major barriers for future power system development, it doesn’t currently pose the fundamental limitations to power conversion technology. It is rather packaging, control, thermal management and system integration issues that are the major barriers limiting the fast growth of power conversion applications. To address these issues, advanced packaging technology is essential. Another important issue for high power density is the packaging of passive components like magnetic and capacitors, which occupied biggest part of the system. With trend of low profile, planar magnetic is a must technology. With

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planar magnetic, passive integration technique could be realized with high power density, low profile and better thermal performance. In this part, the planar metalization device connection, which allows threedimensional integration of power devices, and integration of power passives to increase the power density, as these dominate the physical size of the system, will be discussed.

3.2 Integrated Power Electronics Module for Front end DC/DC Figure 3.3 shows a diagram for a front-end DC/DC converter for distributed power system with asymmetrical half bridge and current doubler configuration.

Figure 3.3 Schematic of front end DC/DC with asymmetrical half bridge current doubler

From previous discussion, we know that the parasitic inductance in the switching commutation path is important to the switching losses and the ringing amplitude. By integrating bare dies of switching devices together using planar integration technology, it can be expected that the parasitic inductance due to packaging be greatly reduced. To further reduce the impact of parasitic inductance, decoupling capacitor is needed. In discrete-component-based design,

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the high frequency decoupling capacitor is paralleled with the DC link electrolytic capacitor and physically installed at the device terminals. By integrating the capacitor into the package, the interconnection from the capacitor to DC/DC switch is simplified and the decoupling effect is improved. Besides the concern of the parasitic inductance in the power path during switching commutation, the common source inductance between gate driver path and the main power path affects switching loss greatly too. During turn-off, the voltage drop on the common source inductance dynamically reduces gate voltage slew rate applied to the gate of MOSFET die, the switching speed is slowed down and this limits the switching loss reduction. Integrating the gate driver circuitry along with the devices further reduces the effective gate driver loop inductance. In the front-end DC/DC converter, the size and dimensions of the passive components strongly affects the total size and volume of the converter. To achieve low profile of the converter, a planar magnetic component is always preferred. To further increase the power density and reduce the profile, there is a demand for the integration of passive components into one planar module, which is known as passive IPEM. The original work of an integrated inductor-capacitor structure led to the development of a planar technology to integrate inductors, capacitors and transformers (L-C-T) for resonant converter applications The integration of passive components for PWM AHB DC/DC converter is another application of L-C-T technology for DC/DC converters.

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Figure 3.4 System partitioning of front end DC/DC converter

From above discussion, the front end DC/DC system could be divided into several blocks as shown in Figure 3.4. One block is the active switches and their drivers that include all the paths sensible to parasitic inductance. Another block is the passive components. With these two blocks, the most critical components for electrical performance and power density are included. Because of the different materials used for these blocks, it is difficult to integrate all these components into one module. Here two IPEMs are been identified and developed: active IPEM and passive IPEM.

3.2.1 Active IPEM For active IPEM, our target is to develop a packaging technology, which provides three-dimensional planar packaging. With three-dimensional planar structure, low profile system could be achieved. Also, with real three-dimensional planar structure, thermal characteristic could be improved. As high switching frequency is our target for system design, lower parasitic is another important criteria for desired packaging technology. Several technologies were been

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developed in CPES. They are Die Dimension Ball Grid Array, Dimple Array Integration, and Embedded Power technology. Flip-chip-on-flex (FCOF) power switching stage modules consist of discrete power chip packages with some of the gate driver components. These chip packages are named Die-Dimensional Ball Grid Arrays (D2BGAs) that comprise a power chip, inner solder bumps, high-lead solder balls, and molding resin. Figure 3.5(a) shows a D2BGA package. The cross section of the package structure is illustrated in Figure 3.5(b). The chip-scale packaged devices are flipping soldered to a patterned flexible substrate. An organic underfill material is introduced into the gap between the packages and substrate to enhance mechanical adhesion and reliability by distributing stresses caused by the mismatched coefficients of thermal expansion between the chip and substrate. To complete the electrical circuitry and achieve good thermal performance, the backsides of the power chips are soldered on to a patterned direct bond copper (DBC) substrate. Finally, the power stage is encapsulated. The photo in Figure 3.5(c) shows a prototype of the packaged FCOF switching power stage module.

Figure 3.5 (a) D2BGA IGBT chip-scale package, (b) schematic of FCOF power switching stage module structure, and (c) FCOF power switching module prototype. 78

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Aiming at enhancing thermal fatigue reliability of solder bumping interconnects in power packaging applications, we have developed the Dimple Array Interconnect (DAI) technique. DAI packaging involves the use of copper flex/sheet with arrays of preformed dimples, which serve as both electrical interconnections and heat removal paths. As shown in Figure 3.6(a), the key feature of DAI is its dimpled metal interconnects, which are convex valleys on metal sheet protruding from one side, that enable easy forming of solder joints with underlying devices. The resultant smooth fillets in solder bumps could significantly reduce thermally induced stresses and strains. An integrated DAI power switching stage module is schematically shown in Figure 3.6(b). The DAI module is realized by solder-attaching DAI power devices, such as diodes and IGBTs, onto DBC substrate, followed by underfilling and encapsulation. Because only one type of solder is needed to form the dimple solder joint, there are more options to select a solder of different melting temperatures for the surface mount (SMT) gate driver and control components. The prototype Dimple Array Interconnect Power DAI power switching is shown in Figure 3.6(c).

Figure 3.6 (a) Schematic of Dimple Array Interconnect, (b) integrated DAI power switching stage module, and (c) prototype Dimple Array Interconnect power switching stage module.

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Embedded power technology has been developed for integrated packaging of integrated power electronics modules (active IPEMs), which are usually comprised of power switches and associated electronics circuitry. Figure 3.7(a) shows the conceptual structure of the embedded power packaged module. It consists of three levels: electronic components, a multiple embedded power chip stage, and a base substrate (from top to bottom). These three parts are soldered together to build a final module. The electronics circuitry includes a gate driver and control and protection components. The base substrate provides electrical interconnection and cooling of power chips. The core element in this structure is the embedded power stage that comprises the ceramic frame, power chips (silicon in the figure), isolation dielectrics and metalized circuit. Inside the power stage, multiple bare power semiconductor dies, featuring vertical semiconductor structures with topside and backside electrode pads, are directly buried in a ceramic frame. Table I summarizes the fabrication steps of an embedded power module. They are the ceramic cutting, device mounting, dielectric printing and metalization. One of the features of this technology is its mask based processing. The metalized base substrate is patterned using photolithography, the dielectric polymer is applied with a screen-printing method, and the chip-carrier ceramic frame is fabricated by computer controlled laser machining.

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(a)

(b)

Figure 3.7 (a) Schematic integration structure of embedded power module and (b) circuit diagram

Compare these three technologies; first two technologies are still based on traditional packaging concept with different interconnection method. Embedded power technology is one step further in three-dimensional packaging, which is a true planar structure with compatible process as planar integration technology. It is used to build the IPEM for distributed power system. Figure 3.7(b) shows the schematic diagram based on the totem-pole structure with high- and low-side drivers and control. The half-bridge consists of two MOSFET devices and gate drivers using three ICs. Figure 3.8 depicts the assembly processes following the embedded power stage and the final packaged module. Figure 3.8(a) and (b) present the top and bottom views of the embedded power stage. Figure 3.8(c) shows components mounted on the topside of the metalized pattern. An Al2O3 DBC substrate with 10mil-thick Cu on both sides of 25 mil-thick ceramic was used as the base substrate, which was pattern etched on one side (Figure 3.8(d)), while the other side is attached directly to the heat spreader. Figure 3.8(e) shows the power stage solder-mounted to the substrate. Finally, after the gate driver had been mounted and connected to the power stage,

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the entire module was encapsulated with protruding input/output pins (Figure 3.8(f)). The completed module measures 30x27x10mm.

Figure 3.8 Assembly process of embedded power module: (a) top view of embedded power stage, (b) back view of embedded power stage, (c) components attachment on top, (d) patterned DBC for base substrate, (e) soldered on substrate, and (f) final encapsulated module.

3.2.2

Passive IPEM

In order to integrate the electromagnetic power passive components into modules, passive integration technology was developed. The technology can be best described by first considering a simple bifilar spiral winding as shown in Figure 3.9. This structure consists of two windings (A-C and B-D), separated by a dielectric material. This resultant structure has distributed inductance and capacitance and is best described as an electromagnetically integrated LC resonant structure for which the equivalent circuit characteristics depend on the external connections. With more winding layers, more complex integrated structures can be realized. This has been demonstrated with an integrated structure 82

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as shown in Figure 3.9. With this technology, the classical term “parasitic” therefore no longer applies and all the higher order impedances are rather referred to as structural impedances. This technology mostly been implemented in resonant converter applications. Here, it is also been used in the asymmetrical half bridge front end DC/DC converter. In this application, the planar passive integration technology, together with planar integrated magnetic technology, were combined to integrated all the high frequency passive components.

Figure 3.9 (a) Spiral integrated LC structure with distributed capacitance and possible external connection configurations, (b) simplified equivalent circuit, and (c) exploded view.

For asymmetrical half bridge converter, two integration steps are used to integrate all the passive components except output filter capacitor into one structure. The schematic of the passive components in asymmetrical half bridge with current doubler is shown in FIG. There are two filter inductor, two transformers, and on DC blocking capacitor. Integrated magnetic concept could be used to

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integrate all the magnetic components into one magnetic structure. This integrated is realized as shown in FIG. It is discussed in detail in [C17][C18][C20]. With two E cores, the two transformers could be constructed on the two outer legs. Its magnetizing inductance is used to build the filter inductors.

Figure 3.10 Magnetic integrated for asymmetrical half bridge converter

As low profile is another important aspect for front end DC/DC converter development, planar magnetic design is preferred to reduce passive components profile. FIG shows the design of planar integrated magnetic. It is built with two planar E cores and one I core. In this structure, I core has the same function as the center leg in previous design.

Figure 3.11 Planar integrated magnetic for current doubler rectifier

With planar integrated magnetic, next the passive integration method will be applied to integrate the DC blocking capacitor into the structure. As shown in FIG, the DC blocking capacitor is implemented in transformer T1 by using the

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hybrid winding technology, The hybrid winding is implemented using copper traces on both sides of the winding and a dielectric layer placed in the middle to enhance the capacitive component of the winding. The transformer T2 is implemented

according

to

conventional

planar

low-profile

transformer

technology. The inductances of the current doubler are realized by the magnetizing inductances of both transformers. Figure 3.12 shows a picture of the final passive IPEM implemented for the AHB DC/DC converter.

Figure 3.12 Explored view and photo of passive IPEM

3.3 Performance evaluation With the use of IPEMs, the front-end converter design takes advantage of the modular design. Besides to the benefit of easy assembly and the reduced overall volume, the active IPEM itself also offers electrical performance improvements, such as the switching loss reduction and the lower voltage stress. Since it is difficult to directly measure the losses in the devices after being installed in the

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converter, the simulation based on SaberTM is used to analyze the effects of the parasitic reduction in the front-end converter. To build the simulation model, the device models for the power devices are first verified. SaberTM provides the modeling tool for the power MOSFETs and some manufacturers provide the model for some devices. However no any manufactures guarantee the accuracy of their models. Therefore efforts must be taken to verify the model accuracy for the expected operating conditions. Since the switching stress and losses are of major concern, the gate charge and the V/I characteristics are obtained in simulation to compare with the data sheet. As shown Figure 3.13 and Figure 3.14, the model matches the data sheet quite well.

Figure 3.13 Gate charge curve of IXFH21N50 (a) Datasheet, (b) simulated

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Figure 3.14 V/I characteristic of IXFH21N50 (a) datasheet and (b) simulated

Next, the parasitic parameters of discrete approach and IPEM are extracted based on the impedance measurement method. The resultant parasitic inductance is shown in Figure 3.15. Compare these two models: first all the parasitic inductances are greatly reduced; second, in active IPEM, the gate loop doesn’t share inductance with power loop. These two aspects will provide significant improvement on electrical performance.

(a)

(b)

Figure 3.15 Parasitic of (a) discrete MOSFET and (b) Active IPEM

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The switching circuit for the active IPEM and the discrete components is built, which includes the parasitic inductances insides the device package and gate driver circuitry. The gate driver pulse is set as ideal square waveform. At 1kW power level, the peak switched current is about 10A. Figure 3.16 shows the simulated waveform of drain source voltage. With active IPEM, the voltage overshoot is reduced to 416V compared with 460V for discrete version. The turnoff loss is reduced from 48uJ to 25uJ by using IPEM. Considering 200kHz operation of DC/DC converter, the turn-off loss is reduced by 8W with 4ohm gate resistance. This translated into about 1% efficiency increase of the front-end converter and 10% loss. Figure 3.17 and Figure 3.18 shows the improvements of active IPEM for different gate resistance.

(a)

(b)

Figure 3.16 Simulation waveforms of Q1 drain source voltage (a) Discrete, and (b) IPEM

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Vds (V)

Figure 3.17 Active IPEM turn-off loss reduction 480 470 460 450 440 430 420 410 400 390

IPEM Discrete

0

2

4

6

8

10 Rg (Ω)

Figure 3.18 Voltage overshoots improvement by IPEM

With passive IPEM, all the passive components except output filter capacitor are integrated into one single package. This could greatly reduce the volume of passive components. Another benefit is from planar design. With planar structure, thermal characteristic of the passive IPEM is improved significantly compared with traditional magnetic structure. In following part, three prototypes were built and compared. They are shown in Figure 3.19. All three prototypes are asymmetrical half bridge with current doubler. First prototype is based on discrete passive design. Second prototype use integrated magnetic technology. Third prototype is built with passive and active IPEM. With discrete passive components, the total volume of passive components is 343cm3. By applying integrated magnetic concept, it is reduced to 258cm3. With passive IPEM, all the passive components except output filter capacitors are integrated. A volume of 87cm3 is achieved, which is five times improvement as shown in Table 3-1.

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(a)

(b)

(c)

Figure 3.19 Photos of three prototypes to be compared (a) discrete design, (b) integrated magnetic design and (c) Passive IPEM design

Figure 3.20 Test efficiency of three prototypes

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Table 3-1 Comparison of three prototypes

(a)

(b)

(c)

Figure 3.21 Temperature test setup for three prototypes Table 3-2 Temperature test results of three prototypes

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Another benefit from passive IPEM is improvements on thermal characteristic. These three prototypes are tested. Figure 3.21 shows the test setup. For each structure, 5 thermal couplers are embedded into the structure. Table 3-2 shows the test results. Compare these three prototypes, one significant improvements of IPEM is the uniform distribution of temperature, which means better thermal characteristic. For discrete version, with more material, although the lowest temperature is lower, hot spot shows even higher temperature than passive IPEM. With integrated magnetic design, the volume is reduced, but hot spot temperature is much higher than discrete version. With passive IPEM, the structure volume is reduced by 5 times; at the same time the temperature is even lower than the discrete version.

3.4 Summary In this chapter, the advanced packaging for front end DC/DC converter is discussed. To achieve high power density and low profile, high switching frequency is necessary. Currently, the power MOSFET could be switches at very high frequency. The high stress and loss limited the ability to use these devices at high switching frequency. These problems are mainly caused by the parasitic components in the circuit. Thermal management also imposes a limitation on the power density achievable with current packaging technology. Two IPEMs are identified for front end DC/DC converter: active IPEM and passive IPEM. Active IPEM mainly focused on the improvements on parasitic

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and thermal issues, while passive IPEM is focused on improved the power density of passive components with advanced passive integration technology. With embedded power technology, two switches in totem pole configuration, gate driver and filter capacitor are integrated with planar 3D packaging. With active IPEM, the parasitic inductance is reduced to less than 10% or discrete devices. With reduced parasitic, voltage stress and switching loss are significantly reduced. Passive IPEM is constructed with combination of different technologies: integrated magnetic, planar magnetic and passive integrate technologies. With passive IPEM, three passive components: transformer, filter inductor and DC blocking capacitor, are integrated into one single package. With passive IPEM, the power density of passive components is improved by 5 times. At the same time, thermal characteristic is significantly improved. The temperature in the passive structure is uniformly distributed. With active IPEM and passive IPEM, the power density and profile of the front end DC/DC converter are improved.

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Chapter 4. LLC Resonant Converter

Chapter 4 LLC Resonant Converter 4.1 Introduction In previous chapters, the trends and technical challenges for front end DC/DC converter were discussed. High power density, high efficiency and high power are the major driving force for this application. Hold up time requirement poses big penalty to the system performance. Two methods were proposed in chapter 2 to solve this problem and improve the efficiency. Range winding solution could improve the performance at high input voltage significantly, but with extra devices, windings and control circuit. Asymmetrical winding solution provides a simpler solution, but could only apply to asymmetrical half bridge topology. Also it introduced other problems like discontinuous output current and unbalanced stress. To catch up with and move ahead of the trend, higher switching frequency, higher efficiency and advanced packaging are the paths we are taking now. Within all these issues, a topology capable of higher switching frequency with higher efficiency is the key to achieve the goal. With the techniques proposed in chapter 2, the performance at normal operation could be improved. But none of these methods dealt with the switching

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loss problem of PWM converter. Even with Zero Voltage Switching technique, the turn on loss could be minimized; turn off loss still limits the capability of the converter to operate at higher switching frequency. Resonant converter, which were been investigated intensively in the 80's [B1][B7], can achieve very low switching loss thus enable resonant topologies to operate at high switching frequency. In resonant topologies, Series Resonant Converter (SRC), Parallel Resonant Converter (PRC) and Series Parallel Resonant Converter (SPRC, also called LCC resonant converter) are the three most popular topologies. The analysis and design of these topologies have been studied thoroughly. In next part, these three topologies will be investigated for front-end application.

4.2 Three traditional resonant topologies In this part, these three topologies will be evaluated for front end DC/DC application. The major goal is to evaluate the performance of the converter with wide input range. For each topology, the switching frequency is designed at around 200kHz.

4.2.1

Series resonant converter

The circuit diagram of a half bridge Series Resonant Converter is shown in Figure 4.1 [B8]-[B13]. The DC characteristic of SRC is shown in Figure 4.2. The resonant inductor Lr and resonant capacitor Cr are in series. They form a series

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resonant tank. The resonant tank will then in series with the load. From this configuration, the resonant tank and the load act as a voltage divider. By changing the frequency of input voltage Va, the impedance of resonant tank will change. This impedance will divide the input voltage with load. Since it is a voltage divider, the DC gain of SRC is always lower than 1. At resonant frequency, the impedance of series resonant tank will be very small; all the input voltage will drop on the load. So for series resonant converter, the maximum gain happens at resonant frequency.

Figure 4.1 Half Bridge Series Resonant Converter

For front end DC/DC application, a SRC is designed to meet the specifications with following parameters: Transformer turns ratio: 5:2, Resonant inductance: 37uH, Resonant capacitance: 17nF.

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Figure 4.2 DC characteristic and operating region of SRC

Figure 4.3 Simulation waveforms of SRC

With above parameters, the range of Q is from 6 (Full load) to 0 (No load). With above design, the operating region of the converter is shown in Figure 4.2 as shaded area. Simulation waveform is shown in Figure 4.3. From the operating region graph and simulation waveforms, several things could be observed:

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Operating region is on the right side of resonant frequency fr. This is because of zero voltage switching (ZVS) is preferred for this converter. When switching frequency is lower than resonant frequency, the converter will work under zero current switching (ZCS) condition. In fact, the rule is when the DC gain slope is negative; the converter is working under zero voltage switching condition. When the DC gain slop is positive, the converter will work under zero current switching condition. For power MOSFET, zero voltage switching is preferred. It can be seen from the operating region that at light load, the switching frequency need to increase to very high to keep output voltage regulated. This is a big problem for SRC. To regulate the output voltage at light load, some other control method has to be added. At 300V input, the converter is working close to resonant frequency. As input voltage increases, the converter is working at higher frequency away from resonant frequency. As frequency increases, the impedance of the resonant tank is increased. This means more and more energy is circulating in the resonant tank instead of transferred to output. From simulation waveforms, at 300V input, the circulating energy is much smaller than 400V input situation. Here the circulating energy is defined as the energy send back to input source in each switching cycle. The more energy is sending back to the source during each switching cycle, the higher the energy needs to be processed by the semiconductors, the higher the conduction loss. 98

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Also from the MOSFET current we can see that the turn off current is much smaller in 300V input. When input voltage increases to 400V, the turn off current is more than 10A, which is around the same level as PWM converter. With above analysis, we can see that SRC is not a good candidate for front end DC/DC converter. The major problems are: light load regulation, high circulating energy and turn off current at high input voltage condition.

4.2.2

Parallel resonant converter

The schematic of parallel resonant converter is shown in Figure 4.4 [B14][B17]. Its DC characteristic is shown in Figure 4.5. For parallel resonant converter, the resonant tank is still in series. It is called parallel resonant converter because in this case the load is in parallel with the resonant capacitor. More accurately, this converter should be called series resonant converter with parallel load. Since transformer primary side is a capacitor, an inductor is added on the secondary side to math the impedance.

Figure 4.4 Half bridge parallel resonant converter

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Figure 4.5 DC characteristic and operating region of PRC

The parameters of parallel resonant converter designed for front end DC/DC application are: Transformer turns ratio: 9:1, Resonant inductance: 58uH, Resonant capacitance: 11.7nF. With above parameters, the range of Q for this converter is 3 (Full load) to ∞ (No load). The operating region of PRC is shown in Figure 4.5 as shaded area. Simulation waveform is shown in Figure 4.6. From the operating region graph and simulation waveforms, several things could be observed:

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Figure 4.6 Simulation waveforms of PRC

1. Similar to SRC, the operating region is also designed on the right hand side of resonant frequency to achieve Zero Voltage Switching. Compare with SRC, the operating region is much smaller. At light load, the frequency doesn't need to change too much to keep output voltage regulated. So light load regulation problem doesn't exist in PRC. Same as SRC for PRC, the converter is working close to resonant frequency at 300V. At high input voltage, the converter is working at higher frequency far away from resonant frequency. From simulation waveforms, at 300V input, the circulating energy is smaller than 400V input situation. Compare with SRC, it can be seen that for PRC, the circulating energy is much larger. Also from the MOSFET current we can see that the turn off current is much smaller in 300V input. When input voltage

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increases to 400V, the turn off current is more than 15A, which is even higher than PWM converter. For PRC, a big problem is the circulating energy is very high even at light load. For PRC, since the load is in parallel with the resonant capacitor, even at no load condition, the input still see a pretty small impedance of the series resonant tank. This will induce pretty high circulating energy even when the load is zero. With above analysis, we can see that PRC is not a good candidate for front end DC/DC converter too. The major problems are: high circulating energy, high turn off current at high input voltage condition.

4.2.3

Series parallel resonant converter

The schematic of series parallel resonant converter is shown in Figure 4.7 [B18]-[B20]. The DC characteristic of SPRC is shown in Figure 4.8. Its resonant tank consists of three resonant components: Lr, Cs and Cp. The resonant tank of SPRC can be looked as the combination of SRC and PRC. Similar as PRC, an output filter inductor is added on secondary side to math the impedance. For SPRC, it combines the good characteristic of PRC and SRC. With load in series with series tank Lr and Cs, the circulating energy is smaller compared with PRC. With the parallel capacitor Cp, SPRC can regulate the output voltage at no load condition. The parameters of SPRC designed for front end DC/DC application are:

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Figure 4.7 Half bridge series parallel resonant converter

Transformer turns ratio: 6:1, Resonant inductance: 72uH, Series resonant capacitor Cs: 17.7nF, Parallel resonant capacitor Cp: 17.7nF, Range of Q: 1 (Full load) to ∞ (No load) The DC characteristic and operating region of SPRC are shown in Figure 4.8. Simulation waveform is shown in Figure 4.9. From the operating region graph, several things could be observed:

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Figure 4.8 DC characteristic and operating region of SPRC

Figure 4.9 Simulation waveforms of SPRC

2. Similar to SRC and PRC, the operating region is also designed on the right hand side of resonant frequency to achieve Zero Voltage Switching.

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From the operating region graph, it can be seen that SPRC narrow switching frequency range with load change compare with SRC. Compare the switching waveforms, the input current in much smaller than PRC and a little larger than SRC. This means for SPRC, the circulating energy is reduced compare with PRC. Same as SRC and PRC,, the converter is working close to resonant frequency at 300V. At high input voltage, the converter is working at higher frequency far away from resonant frequency. Same as PRC and SRC, the circulating energy and turn off current of MOSFET also increase at high input voltage. The turn off current is more than 10A. With above analysis, we can see that SPRC combines the good characteristics of SRC and PRC. Smaller circulating energy and not so sensitive to load change. Unfortunately, SPRC still will see big penalty with wide input range design. With wide input range, the conduction loss and switching loss will increase at high input voltage. The switching loss is similar to that of PWM converter at high input voltage. By analysis, design and simulation of SRC, PRC and SPRC, the conclusion is that these three converters all cannot be optimized at high input voltage. High conduction loss and switching loss will be resulted from wide input range. To

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achieve high switching frequency and higher efficiency, we have to look for some other topologies.

4.3 LLC resonant converter Three traditional resonant topologies were analyzed in above part. From the results, we can see that all of them will see big penalty for wide input range design. High circulating energy and high switching loss will occur at high input voltage. They are not suitable for front end DC/DC application. Although above analysis give us negative results, still we could learn something from it: For a resonant tank, working at its resonant frequency is the most efficient way. This rule applies to SRC and PRC very well. For SPRC, it has two resonant frequencies. Normally, working at its highest resonant frequency will be more efficient. To achieve zero voltage switching, the converter has to work on the negative slope of DC characteristic. From above analysis, LCC resonant converter also could not be optimized for high input voltage. The reason is same as for SRC and PRC; the converter will work at switching frequency far away from resonant frequency at high input voltage. Look at DC characteristic of LCC resonant converter, it can be seen that there are two resonant frequencies. One low resonant frequency determined by 106

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series resonant tank Lr and Cs. One high resonant frequency determined by Lr and equivalent capacitance of Cs and Cp in series. For a resonant converter, it is normally true that the converter could reach high efficiency at resonant frequency. For LCC resonant converter, although it has two resonant frequencies, unfortunately, the lower resonant frequency is in ZCS region. For this application, we are not able to design the converter working at this resonant frequency. Although the lower frequency resonant frequency is not usable, the idea is how to get a resonant frequency at ZVS region. By change the LCC resonant tank to its dual resonant network, this is achievable. As shown in Figure 4.10, by change L to C and C to L, a LLC resonant converter could be built. The DC characteristics of these two converters are shown in Figure 4.11 and Figure 4.12. The DC characteristic of LLC converter is like a flip of DC characteristic of LCC resonant converter. There are still two resonant frequencies. In this case, Lr and Cr determine the higher resonant frequency. The lower resonant frequency is determined by the series inductance of Lm and Lr. Now the higher resonant frequency is in the ZVS region, which means that the converter could be designed to operate around this frequency.

Figure 4.10 LCC and LLC resonant tank

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Figure 4.11 DC characteristic of LCC resonant converter

Figure 4.12 DC characteristic of LLC resonant converter

Figure 4.13 Half bridge LLC resonant converter

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As a matter of fact, LLC resonant converter existed for very long time [B-10] [B-24]. But because of lack of understanding of characteristic of this converter, it was used as a series resonant converter with passive load. Which means it was designed to operate in switching frequency higher than resonant frequency of the series resonant tank of Lr and Cr. When operating in this region, LLC resonant converter acts very similar to SRC. The benefit of LLC resonant converter is narrow switching frequency range with light load and ZVS capability with even no load. In this dissertation, some unexplored operating region of LLC resonant converter will be investigated. Within these operating regions, LLC resonant converter will have some very special characteristic, which makes it an excellent candidate for front end DC/DC application. To design LLC resonant converter, DC analysis is essential. The detailed DC analysis is discussed in Appendix B. Two methods were discussed: fundamental simplification method and simulation method. The error is showed too.

4.4 Operation of LLC resonant converter The DC characteristic of LLC resonant converter could be divided into ZVS region and ZCS region as shown in Figure 4.14. For this converter, there are two resonant frequencies. One is determined by the resonant components Lr and Cr. The other one is determined by Lm, Cr and load condition. As load getting

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heavier, the resonant frequency will shift to higher frequency. The two resonant frequencies are:

fr1 =

1 2 ⋅ π ⋅ Lr ⋅ Cr

fr 2 =

1 2 ⋅ π ⋅ ( Lm + Lr ) ⋅ Cr

With this characteristic, for 400V operation, it could be placed at the resonant frequency of fr1, which is a resonant frequency of series resonant tank of Cr and Lr. While input voltage drops, more gain can be achieved with lower switching frequency. With proper choose of resonant tank, the converter could operate within ZVS region for load and line variation. There are some interesting aspects of this DC characteristic. On the right side of fr1, this converter has same characteristic of SRC. On the left side of fr1, the image of PRC and SRC are fighting to be the dominant. At heavy load, SRC will dominant. When load get lighter, characteristic of PRC will floating to the top. With these interesting characteristics, we could design the converter working at the resonant frequency of SRC to achieve high efficiency. Then we are able to operate the converter at lower than resonant frequency of SRC still get ZVS because of the characteristic of PRC will dominant in that frequency range. From above discussion, the DC characteristic of LLC resonant converter could be also divided into three regions according to different mode of operation 110

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as shown in Figure 4.15. Our designed operating regions are region 1 and region 2. Region 3 is ZCS region. The converter should be prevented from entering region 3. The simulation waveform for region 1 and region 2 are shown in Figure 4.16 and Figure 4.17. In fact, there are many other operating modes for LLC resonant converter as load changes. Those different modes are listed in Appendix B.

Figure 4.14 DC characteristic of LLC resonant converter

Figure 4.15 Three operating regions of LLC resonant converter

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In region 1, the converter works very similar to SRC. In this region, Lm never resonates with resonant capacitor Cr; it is clamped by output voltage and acts as the load of the series resonant tank. With this passive load, LLC resonant converter is able to operate at no load condition without the penalty of very high switching frequency. Also, with passive load Lm, ZVS could be ensured for any load condition. Here the operation will not be discussed in detail. There are several other modes of operation for light load condition. They will be discussed in Appendix B.

Figure 4.16 Simulated operation waveforms in region 1

In region 2, the operation of LLC resonant converter is more complex and interesting. The waveforms could be divided into clearly two time intervals. In

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first time interval, Lr resonant with Cr. Lm is clamped by output voltage. When Lr current resonant back to same level as Lm current, the resonant of Lr and Cr is stopped, instead, now Lm will participate into the resonant and the second time interval begins. During this time interval, the resonant components will change to Cr and Lm in series with Lr, which is shown in the waveforms as a flat region. In fact, that is a part of the resonant process between Lm+Lr with Cr. From this aspect, LLC resonant converter is a multi resonant converter since the resonant frequency at different time interval is different. Because of the resonant between Lm and Cr, a peak on the gain appears at resonant frequency of Lm+Lr and Cr. Next the operating of LLC resonant converter in region 2 will be discussed in detail. It is divided into three modes.

Figure 4.17 Simulation waveforms in region 2

Mode 1 (t0 to t1):

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This mode begins when Q2 is turned off at t0. At this moment, resonant inductor Lr current is negative; it will flow through body diode of Q1, which creates a ZVS condition for Q1. Gate signal of Q1 should be applied during this mode.

Figure 4.18 Circuit diagram during mode 1 in region 2

When resonant inductor Lr current flow through body diode of Q1, ILr begins to rise, this will force secondary diode D1 conduct and Io begin to increase. Also, from this moment, transformer sees output voltage on the secondary side. Lm is charged with constant voltage. Mode 2 (t1 to t2) This mode begins when resonant inductor current ILr becomes positive. Since Q1 is turned on during mode 1, current will flow through MOSFET Q1.

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Figure 4.19 Circuit diagram during mode 2 in region 2

During this mode, output rectifier diode D1 conduct. The transformer voltage is clamped at Vo. Lm is linearly charged with output voltage, so it doesn't participate in the resonant during this period. In this mode, the circuit works like a SRC with resonant inductor Lr and resonant capacitor Cr. This mode ends when Lr current is the same as Lm current. Output current reach zero. Mode 3 (t2 to t3) At t2, the two inductor’s currents are equal. Output current reach zero. Both output rectifier diodes D1 and D2 is reverse biased. Transformer secondary voltage is lower than output voltage. Output is separated from transformer.

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Figure 4.20 Circuit diagram during mode 3 in region 2

During this period, since output is separated from primary, Lm is freed to participate resonant. It will form a resonant tank of Lm in series with Lr resonant with Cr. This mode ends when Q1 is turned off. As can be seen from the waveform, Q1 turn off current at t3 is small compare with peak current. For next half cycle, the operation is same as analyzed above. From the simulation waveform we can see, the MOSFETs are turned on with ZVS. The ZVS is achieved with magnetizing current, which is not related to load current, so ZVS could be realized even with zero-load. Since this magnetizing current is also the turn off current of MOSFET. Choosing different magnetizing inductance could control it. The turn off current could be much smaller than load current, so turn off loss can be reduce. Also, the secondary side diode current reduce to zero and stay off, the reverse recovery is eliminated also. With all these, the switching loss of this converter is very small.

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4.5 Design of LLC resonant converter 4.5.1 Design of power stage parameters

From above analysis, the DC characteristic of LLC resonant converter could be derived. Based on the DC characteristic, parameters in power stage can be designed. The parameters need to be designed are: •

Transformer turns ratio: n

•

Series resonant inductor: Lr

•

Resonant capacitor: Cr

•

Resonant inductor ratio: Lm/Lr The specifications for the design are: Input voltage range: 300V to 400V, normal operating region (360-400V) Output voltage: 48V Maximum load: 2.5Ohm Maximum switching frequency: 200kHz With above information, we can begin to choose the parameters. For front-end application, the target is to optimize the performance at high

input voltage. From previous analysis results, the optimal operating point for this

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converter is when switching frequency equals to the resonant frequency of Lr and Cr. At this point, the voltage gain of LLC resonant converter is 1. Base on this, the transformer turns ratio can be choose. For Half Bridge LLC resonant converter with 400V input and 48V output, the transformer turns ratio can be choose base on following equations: n = Vin /( 2 ⋅ Vo)

For Full Bridge LLC resonant converter, the turn’s ratio will be: n = Vin / Vo

In our design, a half bridge LLC resonant converter is used; the turns ratio was choose to be 4. After the transformer turns ratio, the resonant tank can be designed. To determine the resonant tank, lot of trade offs are involved. Three design examples will be shown to demonstrate the trade offs. Design 1: In this design, the ratio of two resonant inductors is 1, which means the two resonant inductors are with same value. The characteristic and operating region are shown in Figure 4.21. The region of Q is from 1(Full load) to 0(no load). Here Q is defined as: Q=Zo/Rl. Resonant inductor Lr is 27.8uH, resonant inductor Lm is 27.8uH and Cr is 22.8nF.

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Figure 4.21 Operating region for design 1

Simulation waveform is shown in Figure 4.22.

Figure 4.22 Simulation waveforms of design 1 with 300V and 400V input voltage

Design 2: In this design, the ratio of two resonant inductors is 4, which means Lm is four times Lr. The characteristic and operating region are shown in Figure 4.23.

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The region of Q is from 0.5 (Full load) to 0 (no load). Resonant inductor Lr is 17uH and Lm is 70uH. Resonant capacitor is 24nF.

Figure 4.23 Operating region for design 2

Simulation waveform is shown in Figure 4.24.

Figure 4.24 Simulation waveforms of design 2 with 300V and 400V input voltage

Design 3:

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In this design, the ratio of two resonant inductors is 16, which means Lm is sixteen times Lr. The characteristic and operating region are shown in Figure 4.25. The region of Q is from 0.25 (Full load) to 0 (no load). Resonant inductor Lr is 7.95uH, Lm is 127uH and Cr is 79.6nF.

Figure 4.25 Operating region for design 3

Simulation waveform is shown in Figure 4.26.

Figure 4.26 Simulation waveforms of design 3 with 300V and 400V input voltage

Summary of three designs: 121

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Table 4-1 Summary of three LLC resonant converter designs

Design 1 Design 2 Design 3

Q range

Fs range

1 to 0 0.5 to 0 0.25 to 0

175k to 200k 135k to 200k 72k to 200k

Primary RMS Current 8.1A to 9.2A 6.0A to 8.3A 5.7A to 10.2A

Switch turn off Resonant Cap current Voltage 7.8A to 5.8A 800V 4.1A to 3.2A 440V 1.9A to 0.24A 430V

Peak Output current 31A to 43A 31A to 49A 31A to 89A

From the summary, design 3 provides best performance at 400V input, but the switching frequency range will be much larger. For design 1, the performance at 400V is compromised; the benefit is very narrow switching frequency range. For this application, since the output voltage of PFC circuit is not tightly regulated, it has a range from 360 to 400V. The performance at 360V is also a concern. In Figure 4.27 to Figure 4.29, primary switching and conduction loss are compared for three designs with different input voltage.

Figure 4.27 Primary loss for three designs with Vin=400V

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Figure 4.28 Primary loss for three designs with Vin=360V

Figure 4.29 Primary loss for three designs with Vin=300V

From these comparisons, although design 3 could provide better performance at 400V input, its performance degrades very fast as input voltage drops. Design 1 could provide more balanced performance for whole range, but the performance at 400V input is greatly impaired. Design 2 is choosing for front-end application with 200kHz design. With design 2, the performance is balanced within input range. Stress on different devices is reasonable.

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4.5.2 Design trade offs:

In design of power stage, there are some trade offs that will affect the final results. First trade off is switching frequency range and switching loss. With smaller magnetizing inductance, narrower switching frequency range can be achieved, but switching loss and conduction loss will increase because of high magnetizing current. Another trade off to make is switching frequency range and resonant tank impedance. For same specification, Lr and Cr can have different values, which will work. Although there is a limit on how small Cr can be in order to keep series resonant tank work in constant gain region. With larger Cr, the voltage stress on Cr will be smaller. The problem is that the impedance of the resonant tank will be small too, which will affect the short circuit performance. With smaller tank impedance, the higher the shorts circuit current will be and higher switching frequency is needed to limit the output current. The problem with low switching frequency is the conduction loss will increase as switching frequency drops. As shown in first part, the conduction loss can be doubled when switching frequency change from 200kHz to 150kHz. From these trade offs, the optimized design should choose as small resonant capacitor as possible to get enough voltage gain at heavy load. Then Lm should

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be as large as possible to get the voltage gain with desired switching frequency range.

4.5.3

Test results

Base on the design, a LLC resonant converter is built with those parameters. The test circuits are shown in Figure 4.30 with the part number of the devices. Test waveforms are shown in Figure 4.31. The test efficiency is shown in Figure 4.32. Compare with asymmetrical half bridge converter, LLC resonant converter could improve the efficiency at normal operation point by more than 3%. Figure 4.33 shows the test efficiency at different input voltage. LLC resonant converter could cover wide input range with much higher efficiency compared with PWM converter.

(b)

Figure 4.30 Test circuit for 200kHz LLC resonant converter

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(a)

(b)

Figure 4.31 Test waveform of LLC converter at full load and (a) 300V input, and (b) 400V input

Figure 4.32 Test efficiency of LLC converter and HB converter at 400V input

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Figure 4.33 Test efficiency of LLC and AHB converter at different input voltage and full load

4.5.4

Loss comparison

Since our target is to achieve high efficiency, and high frequency. In this part, the loss of LLC resonant converter is compared with Asymmetrical Half Bridge and asymmetrical winding asymmetrical half bridge. Comparison is based on simulations and calculations. All converters are designed with same specifications: Vin=300 to 400V, Vout=48V, and Pout=1kW; Switching frequency for PWM 200kHz and switching frequency for LLC is from 140kHz to 200kHz. The design of Asymmetrical half bridge just follows the traditional design procedure. LLC resonant is as shown in Figure 4.30. The circuit diagrams of both converters with devices listed are shown in Figure 4.34.

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(a)

(b)

Figure 4.34 Two PWM topologies to be compared (a) AHB, and (b) asymmetrical winding AHB

The devices we used for these converters are: Primary switches for both converters: IXFN24N50 500V 21A MOSFET Asymmetrical Half Bridge secondary rectifier: STTH3003, 300V ultra fast diode, forward voltage drop 1.2V. Asymmetrical winding AHB: MBR20200, 200V shottky diodes with forward voltage drop 0.95V. LLC resonant converter secondary rectifier: IR30CPQ150, 150V shottky diodes with forward voltage drop 0.65V.

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For PWM Half Bridge converter, the current waveform is highly asymmetrical at high input voltage. This will increase both the conduction loss and switching loss when the converter works in this condition. So the efficiency of the converter will be hurt by wide input range. The small duty cycle will also increase the voltage stress of the secondary rectifier. Higher voltage rated devices have to be used which have higher forward voltage drop. Secondary conducting loss will be a large part of total loss. For LLC resonant converter, at high input voltage, the input current have lower peak value and RMS value, so the conduction loss is much lower at high input voltage. Also, the secondary side voltage stress is fixed at two times output voltage for LLC resonant converter. Low voltage schottky diodes can be used to reduce the secondary conduction loss. For resonant converter, high current stress or voltage stress is always a concern since the increase of conduction loss will annihilate the benefit get from reduced switching loss.

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Figure 4.35 Primary side conduction loss comparison

Figure 4.35 shows the primary conduction loss comparison. Because for PWM converter, at high input voltage, the duty cycle is small, so the RMS current is even higher than LLC resonant converter. When the input voltage decrease, the duty will become more symmetrical and the resonant converter will show higher RMS current and higher conduction loss. But this is not a problem for this case since our normal working condition input voltage will be within 360V to 400V. Only during fault condition would the circuit work at so low input voltage. With LLC resonant converter, primary conduction loss is not a problem. Figure 4.36 shows the primary switching loss comparison of Half Bridge and LLC resonant converter based on the datasheet. It can be seen that switching loss of LLC is 40% lower than PWM converter. This calculation is for 200kHz switching frequency, if we increase the switching frequency further, this will make even bigger difference.

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Figure 4.36 Primary switches turn off loss comparison

Another benefit of LLC resonant converter is that the voltage stress on the secondary rectifier is much lower than asymmetrical half bridge. For Asymmetrical half bridge converter, the highest voltage stress on secondary diode will be higher than 250V; 300V diode has to be used. With asymmetrical winding, the voltage stress could be reduced so that 200V diodes could be used, still the forward voltage drop is pretty high at 0.95V. For LLC converter, the voltage stress is limited to two times output voltage, so 150V shottky diode can be used. In this comparison, STTH3003, which has forward voltage drop of 1.2V, is used for PWM converter. For LLC converter, we choose 30CPQ150, which has forward voltage drop of 0.65V. This will reduce the secondary conduction loss from 25W to 13W. This means 1% efficiency improvement.

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Figure 4.37 Secondary conduction loss comparison

There is still something not included into this analysis: the rectifier diodes reverse recovery loss. In PWM converter, the diodes commutate at load current with high dv/dt; this will cause pretty high reverse recovery loss. In LLC resonant converter, since the diode current resonant to zero, this part of loss is eliminated. From the test results, there is no ringing on the secondary diodes without any snubber circuits. Table 4-2 Loss breakdown comparison of AHB and LLC

Table 4-2 shows the loss breakdown for asymmetrical half bridge and LLC resonant converter. Without considering secondary reverse recovery loss, LLC

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resonant converter can provide about 2% efficiency improvement over asymmetrical half bridge. For both topologies, we can achieve zero voltage turn on, so we assume turnon loss could be neglected. In fact, for PWM converter, we are using the energy stored in leakage inductance to achieve ZVS, so at light load, the converter will loss ZVS capability. But for LLC resonant converter, we are using magnetizing inductor to achieve ZVS, so we are able to get ZVS in whole load range. From previous discussion we can see that with LLC resonant converter, we can get high efficiency and high switching frequency, which is the key to meet the trend for this application.

4.6 Extension of LLC resonant topology As demonstrated in previous part, the benefit of LLC resonant converter comes from the fact that it has two resonant frequencies and the operating point of normal operating condition could be positioned at resonant frequency. In three elements resonant tanks, there are 36 different configurations. The question is if there is any other resonant tank configuration could provide similar characteristic so that could result to similar benefits. To answer this question, a thorough search is performed for three resonant components resonant tank as shown in Appendix A. Since our application is a voltage source input application. Only resonant tank could be used for voltage source input are look in detail,

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which consists 23 resonant tank configurations. With in these 23 resonant tank configurations, three resonant tanks show the desired characteristic as shown in Figure 4.38. Tank G is the same resonant tank as we used for LLC resonant converter. It is interesting to notice that the other two resonant tanks also have two resonant inductors and one resonant capacitor.

Figure 4.38 Resonant tanks with desired DC characteristic

For resonant tank W, a resonant converter could be constructed as shown in Figure 4.39. With this converter, the series resonant inductor will be on secondary side. This topology operates same as LLC resonant converter discussed before except the voltage stress on the output rectifier diodes will be different as will be discussed in magnetic design in next chapter. With resonant tank U, a half bridge resonant topology could be constructed as shown in Figure 4.40. These three topologies have similar characteristic. They will be able to operate at resonant frequency when input voltage is high while still cover wide input range.

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Figure 4.39 Half bridge resonant converter with tank W

Figure 4.40 Half bridge resonant converter with tank U

4.7 High frequency operation LLC resonant converter shows much lower switching loss compared with PWM converter as shown in previous part. Compare with PWM converter, the primary switching loss is reduced by more than 50%. Secondary diode commute naturally so there is no reverse recovery problem at all. All these enable LLC resonant converter to be able to operate with higher switching frequency. Another significant difference of LLC resonant converter and PWM converter is that now switching loss could be controlled. For PWM converter, switching loss is determined by device and load condition. To deliver the power, the switches have to turn off significant amount 135

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of current. For asymmetrical half bridge, it is around 13A of current. This will introduce very high switching loss. This is also one limitation why PWM converter could not operate with higher switching frequency. For LLC resonant converter, switching loss is a controlled parameter. As shown in Figure 4.41, the turn off current of primary switching is determined by the choice of resonant inductor Lm. With larger Lm, turn off current will be smaller, vice versa. This could also be observed from the design examples shown in previous section.

Figure 4.41 Simulation waveform of LLC resonant converter

In this part, a 400kHz LLC resonant converter is designed to demonstrate the benefits of LLC resonant converter with high switching frequency. The resonant tank is scale down as designed for 200kHz LLC resonant converter. For design of Lm, balance of switching loss and switching frequency range is considered. As shown in Figure 4.42, with smaller Lm, switching loss will increase. In this case, 40uH Lm is chosen which gives less than 10W switching loss. 136

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Figure 4.42 Switching loss with different Lm for 400kHz LLC resonant converter

Figure 4.43 Circuit diagram of 400kHz LLC resonant converter

The magnetic design use integrated magnetic concept, which will be discussed in next chapter.

Figure 4.44 Integrated magnetic structure for 400kHz LLC resonant converter

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The parameters are as following: Turns ratio: 12:3; Lr: 7uH; Lm: 40uH. Design results are: np1 = 6, np2 = 10, ns = 3 and gap = 0.5mm. The magnetic components for 200kHz LLC and 400kHz LLC resonant converter are shown in Figure 4.45. Both magnetic components use integrated magnetic concept, which will be discussed in next chapter. With 400kHz switching frequency, volume of the magnetic component is reduced by more than 40%. The test circuits are shown in Figure 4.46. Compare 200kHz LLC resonant converter with 200kHz AHB, the power density could be improved by almost 100%. With high frequency, the power density is improved by 200%.

(a)

(b)

Figure 4.45 Magnetic components size comparison (a) 200kHz and (b) 400kHz

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(a)

(b)

(c)

Figure 4.46 Test circuits of front end DC/DC converter (a) 200kHz AHB, (b) 200kHz LLC, and (c) 400kHz LLC

Figure 4.47 shows the test efficiency. With 400kHz switching frequency, the efficiency is lower than 200kHz. But compare 400kHz LLC resonant converter with 200kHz PWM converter, the efficiency of LLC resonant converter is still better. The projected 400kHz PWM converter uses the test data of 200kHz PMW converter with consideration of switching loss of primary switches at 400kHz. In

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fact, the efficiency will be even lower with consideration of secondary diode reverse recovery loss.

Figure 4.47 Test efficiency of 400kHz LLC resonant converter

4.8 Summary In this chapter, resonant topologies were investigated for front end DC/DC application. The target is to find a topology, which could be optimized at high input voltage with low switching loss. First three traditional resonant topologies were designed and simulated for this application. They are: series resonant converter, parallel resonant converter and series parallel resonant converter. Unfortunately, all three converters suffered from wide input range problem and could not be optimized for high input voltage. LLC resonant topology was introduced by observing the tank characteristic of series parallel resonant converter. With LLC resonant converter, performance at high input voltage could be optimized and the converter still could cover wide input voltage range. Test results shows that with LLC resonant converter, the

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efficiency could be improved by more than 3% compared with PWM converter. With high switching frequency, LLC resonant converter could achieve 45W/in3 compared with 13W/in3 of PWM converter. From analysis and test, LLC resonant converter is proved to be able to improve the performance of front end DC/DC converter significantly. Another question is that if this is the only resonant tank or there is more resonant tank, which could provide similar characteristic.

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Chapter 5 Improvements of LLC Resonant Converter From previous chapter, the characteristic and design of LLC resonant converter were discussed. In this chapter, two improvements for LLC resonant converter will be investigated: integrated magnetic design and over load protection.

5.1 Magnetic design for LLC Resonant Converter From previous discussion, the power stage could be designed according to the given specifications. The outcome of the design is the desired values for the components. For these components, power devices and capacitors are obtained from manufactures, which already reflect the state of the art technology. Within all these components, magnetic is the one need to be physically designed and built by power electronics researcher. In this part, the design of magnetic component for LLC resonant converter will be discussed.

5.1.1 Discrete design and issues For a LLC resonant converter, the magnetic components need to be designed are shown in Figure 5.1. There are three magnetic components: Lr, Lm and transformer T. From the configuration of Lm and transformer T, it is easy to build

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Lm as the magnetizing inductance of transformer. So in fact, we are trying to build one resonant inductor and one transformer with magnetizing inductance.

Figure 5.1 Magnetic structure for LLC resonant converter

There are several ways to build them. One is using discrete components, with one magnetic core to build the resonant inductor and one magnetic core to build the transformer and magnetizing inductor Lm. The benefit of this method is that the design procedure is well established. Next, a discrete design is presented and simulation result is showed to provide a reference for later integrated magnetic designs. For LLC resonant converter, the resonant inductor Lr has pure AC current through it, so we use soft ferrite core for both inductor and transformer. Figure 5.2 shows the discrete design of the magnetic for LLC resonant converter. Two U cores were used to build the resonant inductor and gapped transformer. Fig.6 shows the simulation results of flux density in the core. For

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each U core, the cross-section area is 116.5mm2. Design result: nl=12, np: ns: ns=16:4:4, gap1=1.45mm and gap2=0.6mm.

(a)

(b)

Figure 5.2 Discrete magnetic design (a) schematic (b) physical structure

(a) Inductor

(b) Transformer Figure 5.3 Flux density simulation result (a) Inductor, and (b) Transformer

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Figure 5.3 shows the flux density in each core at 400V input with switching frequency at 200kHz. As seen in the graph, the flux densities in both cores are pretty high. Both cores with high flux density excitation will contribute to the total core loss. For high frequency, core loss is a major limitation on pushing to higher frequency and smaller size. Figure 5.4 shows the peak-to-peak flux density for each core with different input voltage. At low input voltage, the flux density will increase, but it is not critical because of short operating time.

(a) Inductor

(b) Transformer Figure 5.4 Peak to peak flux density under different input voltage at full load 145

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The drawbacks of this method are: 1. Two magnetic cores are needed, which results in more components count and connections, 2. High magnetic loss caused by high flux ripple in magnetic structure, 3. Large footprint is needed for the whole structure. In recent years, integrated magnetic has been investigated for many different applications. For asymmetrical half bridge with current doubler, all the magnetic components could be integrated into one magnetic structure with integrated magnetic concept [C1][C5]. In this part, the integrated magnetic structure will be discussed for LLC resonant converter. It integrated all magnetic components into one magnetic core. Through magnetic integration, the component count and footprint are reduced, the connections is also reduced. With proper design; flux ripple cancellation can be achieved, which can reduced the magnetic loss, and reduce the magnetic core size. In the next part, the integrated magnetic designs for LLC resonant converter will be discussed and compared.

5.1.2 Integrated magnetic design 5.1.2.1 Real transformer with leakage and magnetizing inductance First structure is just use one transformer and uses the leakage inductance as resonant inductor. The configuration of magnetic components for LLC resonant converter is exactly the same as a real transformer with magnetizing inductance

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and leakage inductance. It is natural to think about using one real transformer to get all the needed components. The issues with structure are: 1. The leakage inductance cannot be accurately controlled which will determine the operating point of the converter, 2. When we build Lr this way, the leakage inductance will not only exist on primary side, it will also exist on secondary side of the transformer. So the result get from real transformer will be as in Figure 5.5. Llp and Lls have similar value when transferred to same side of the transformer.

Figure 5.5 Structure with real transformer

Figure 5.6 Desired magnetic components configuration

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Figure 5.7 Magnetic components configuration from real transformer

(a)

(b) Figure 5.8 Voltage stress of output diodes D1 D2 (a): desired structure (b) real transformer

When the leakage inductance exists on secondary side, it will increase the voltage stress on secondary rectifier diode. This requires us to use higher voltage rating diode, which will increase the conduction loss of the output rectifier. Figure 5.8 shows the simulate waveforms of secondary diodes voltage stress with 148

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magnetic structure in Figure 5.6 and Figure 5.7. We can see that with inductor on the secondary side, the voltage stress of the diodes is much higher. From above discussion, we can see that the desired magnetic structure will need to provide accurate control of Lr and Lm, at that same time, minimize the inductance on secondary side, which could not be achieved with just a transformer with leakage and magnetizing inductance. Next more complex integrated magnetic structure will be investigated.

5.1.2.2 Integrated magnetic design A From discrete design, just combine them together with an EE core, we will be able to integrate the two components into one magnetic component as shown in Figure 5.9.

Figure 5.9 Integrated Magnetic Designs A

E42/21/20 core is used. The cross-section area of is 233mm2. For the outer legs, they have same cross-section area as discrete design. Turn number nl, np and ns is the same as in discrete design. For this design, the inductor and transformer

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design is decoupled. Discrete design procedure still can be used. Figure 5.10 shows the simulation result of for this structure.

Figure 5.10 Flux density simulation result for Design A

It can be seen from the simulation result: for inductor and transformer leg, the flux density is the same as discrete design. But for center leg, the flux density is much smaller than discrete case. This will greatly reduce the magnetic loss in the big part of the magnetic component.

Figure 5.11 Center leg flux density for different input voltage

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Figure 5.11 shows the center leg flux density for whole input voltage range. Compare with discrete design, the flux density is only half of the transformer leg and much smaller than inductor leg within all input voltage range. The problem for this structure is the gapping. In this structure, we are using E cores. The air gap is on two outer legs while there is no air gap on center leg. This structure is not good in several aspects: first, this core structure is not a standard. The standard core normally has air gap on the center leg or no air gap at all. Second, it is not a mechanical stable structure, very accurate gap filling need to be provided. Otherwise, the accuracy of the components value will be impacted. Also, when force is applied which happens when the converter is working, the core tends to vibrate. This vibration will cause broken of the core. A desired core structure will have air gap on center leg or same air gap for all three legs. Following part will try to establish an electrical circuit model for a general integrate magnetic structure. From the model, we can investigate new core structures.

5.1.2.3 Extraction of Common Structure for Integrated Magnetic In the past, lot of research was done on integrated magnetic design for power converters. Review those paper, we can find that most of them are based on EE core structure or three legs structure. The difference between different designs is the placement of windings and air gaps.

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In this part of the paper, the general circuit model of an EE core with four windings is used as a general structure as shown in Figure 5.12. There are air gaps on each leg. This is a very commonly used structure, many integrated magnetic design for PWM converter also used this structure with some change on the air gap or winding placement [C5]. The reason of choosing this structure for LLC resonant converter is as following: To integrate two magnetic components, usually we need three magnetic paths. In the LLC resonant converter, although we have three magnetic components, Lm and transformer T can be build with an air-gapped transformer. So in fact we need integrated two magnetic components: series resonant inductor Lr and gapped transformer T. An EE core structure will be a reasonable choice.

Figure 5.12 general magnetic structures for Integrated magnetic

The model is derived through duality modeling method [E4]. Through this method, we can get the electrical circuit model of a physical magnetic structure. All the components in the model are related to the physical structure of the magnetic structure. Figure 5.13 shows the reluctance model of magnetic structure

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shown in Figure 5.12. Figure 5.14shows electrical circuit model form this structure. In the structure, we have two sets of ideal transformer and three inductors.

Figure 5.13 Reluctance model of general integrated magnetic structure

For the two ideal transformers, they have same turns ratio as in real physical structure. For the three inductors, they are correspond to each air gap and reflected to first winding n1. They can also be reflected to other windings as necessary. The value of each inductors are as following:

Figure 5.14 Circuit model of general integrated magnetic structure

Base on this circuit model, we will investigate more integrated magnetic structures.

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5.1.2.4 Integrated magnetic design B for LLC resonant converter As discussed in structure A, the air gapping for structure A is not easy to implement. In this part, we will investigate structure with same air gap for all three legs and same winding structure as shown in Figure 5.15.

Figure 5.15 Integrated Magnetic Designs B

The electrical model of this structure can be easily got from general structure. Compare this structure with general structure; design B has only one winding on left side leg. By simplify the general model we can get following circuit model of design B as shown in Figure 5.16.

Figure 5.16 Electrical circuit model of integrated magnetic structure B

Base on the electrical circuit model of the structure, next terminal 2 and 3 are connected, which gives following circuit model.

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Figure 5.17 Electrical model of connecting dot-marked terminal with unmarked terminal

From circuit model in Figure 5.17, write the input current and voltage equations and solve them, then we can get the equivalent circuit of the structure. For this circuit, it has two modes. One mode is n3 is connected to output voltage. During this mode, the energy is transferred from primary to output. During the other mode, both secondary windings n3 are not connecting. We will derive the equivalent circuit for these two modes separately.

(mode a)

(mode b) Figure 5.18 Two operation modes for LLC resonant converter

For operation mode (a), we can get following equations:

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L1

di1 n2 + v1 = vin dt n1

L0

di0 n2 + v1 + v1 = vin dt n1

v1 =

n1 Vo n3

i0 + i1 = iin From above equations, we can get the relationship of input voltage, input current and output voltage as following:

vin =

L1 ⋅ L0 diin 1 L1 + Vo (n2 + n1 ) L1 + L0 dt n3 L1 + L0

From this equation, we can get the equivalent circuit during this mode as in Figure 5.19.

Figure 5.19 Equivalent-circuit for mode (a)

In this circuit, Lr, Lm and na are as following: Lr =

L1 ⋅ L0 L1 + L0

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na = n2 + n1

L1 L1 + L0

To find out Lm, we need to analyze mode (b). Same as analysis for mode (a), we can get following equations for Lm:

Lm = L2 ⋅

na 2 L1 + L0 ⋅ 2 n1 L1 + L2 + L0

From the equivalent circuit, derive the relationship between terminals; the equivalent circuit above can be simplified into the equivalent circuit, which is the structure we desired. The relationship between resonant inductor, magnetizing inductance and transformer turns ratio is shown also. Base on these equations, the structure can be designed. Following is an example of design: turns ratio 12:3, Lm=14u and Lm = 60uH. The relationship of above equations could be drawn in Figure 5.20. For given turns ratio, there are many different ways to choose n1 and n2 to get the desired na, for example, n1=n2=9, n1=6 and n2=10. The other constrain will be the desired Lm. For this case, the Lm is 4.5 times Lr. To get this Lm, the n2 need to be choosing as 10.

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(a)

(b)

Figure 5.20 Design curves for integrated magnetic structure B for LLC converter

From above discussion, n1=6, n2=10 and n3=3 give us turns ratio 12:3, Lm/Lr = 4.5. Next step will be design the air gap, we knows n1 and L1 value. Follow tradition inductor design equations, the air gap can be designed. Here Lr = 14uH, from the structure it can be seen that: L1 = L3 = 0.5 L2. From the relationship above, it can be calculated that we need L1=21uH to give us equivalent Lr=14uH. With the core cross-section area and turns given, the gap can be easily derived. In this part, the detailed information of the magnetic is described. For this converter, the core used is EE56/24/19 from Phillips. The core material is 3F3. Two outer legs are used to wind the windings. Air gap is 0.55mm for all legs. Primary windings are built with 8 strands of AWG#27 wires. Secondary side uses 5mil X 0.9inch copper foil.

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Figure 5.21 shows the simulated flux density on each of the legs. From simulation result we can see that the flux density on center leg is greatly reduced. So with this integrated magnetic structure, we can reduce the core loss greatly. Also, with this structure, the air gap is the same for all legs, which is easier to manufacture and doesn’t have mechanical problem.

Figure 5.21 Flux density in each leg for integrated magnetic structure B

5.1.3 Test Result In this part, the test result of integrated magnetic structure B is tested. It is compared with a discrete design. The test efficiency of integrated magnetic and discrete magnetic is shown in Figure 5.22. Because of flux ripple cancellation effect and less turns number, although the size of the magnetic components is reduced, the efficiency is almost the same for these two designs. In Figure 5.23, the sizes of these two designs were compared. With integrated magnetic, the footprint of the magnetic components could be reduced by almost 30%.

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Figure 5.22 Efficiency comparison of integrated and discrete magnetic design for LLC converter

Figure 5.23 Magnetic size comparison of discrete and integrated magnetic

5.1.4

Summary

In this part, the magnetic design for LLC resonant converter is discussed. Discrete design and three method of integrated design were investigated. For discrete design, the footprint is pretty large. Also, there is no flux ripple cancellation effect; the magnetic loss is high in discrete design too. With real transformer, the magnetic components could be built with one magnetic structure. The problem is difficult to control the leakage inductance. Another integrated

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magnetic structure is to integrate the two U cores used to build discrete magnetic. With this method, the problem is the mechanical structure is not a stable structure. To improve this structure, a general integrated magnetic structure is developed. With the model, another integrated magnetic structure is developed with same air gap on all legs. With this magnetic structure, the manufacture is easy. There is no mechanical problem. Also, flux ripple cancellation could be achieved with this structure. Compare with discrete design, the integrated magnetic structure could provide same efficiency with 30% less footprint.

5.2 Over load protection for LLC resonant converter In previous part of this chapter, the design of power stage was discussed. Base on these discussions, the power stage of LLC resonant converter could be designed for given specifications. Magnetic design is also investigated for LLC resonant converter. Till now we got a converter could convert 400V DC to 48V DC output with high efficiency and high power density. However, to make practical use of this converter, there are still some issues to be solved. Over load protection is one critical issue, which will be discussed in this part. The purpose of over load protection is to limit the stress in the system during over load condition. Another function is to limit the inrush current during start up when output voltage is zero so that the power converter can be protected from destructive damage under those conditions.

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In some applications, continuous operation in over load condition is required in order to achieve high system availability. In order to achieve this target, other than limit the current, healthy operation is also an important consideration, which means when the converter is running into over load protection mode, the operation of semiconductor and other components should not cause destructive damage too, i.e. lost of Zero Voltage Switching, body diode reverse recovery etc. For traditional PWM converter, during over load condition, duty cycle is reduced to limit the current. With smaller duty cycle, the current stress could be limited. For LLC resonant converter, it is working with variable frequency control at constant 50% duty cycle. The over load protection is totally different story. To investigate the over load protection method for LLC resonant converter, following questions need to be answered. First, the intrinsic response of LLC resonant converter to over load situation needs to be understood. Second, methods to improve the intrinsic response need to be developed if the intrinsic response is not safe or healthy. In this part of the dissertation, first the intrinsic response of LLC resonant converter to over load condition will be investigated. Then three different over load protection methods will be discussed. First method is increasing the switching frequency. The second method, a combination of variable-frequencycontrol and PWM control is used to achieve over load protection. In the last 162

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method, the power stage is modified to include current limiting function into the converter. In following sections, each method will be discussed separately. The parameters for the LLC converter used in this discussion are: Lr=12uH, Cr=33nF and Lm=60uH, transformer turns ratio: 4:1. With above specs and parameters, the switching frequency range for the converter is: 170kHz to 250kHz.

5.2.1 Intrinsic response of LLC resonant converter to over load condition In LLC resonant converter, the impedance of the resonant tank is pretty low during normal operation because it is working close to the resonant frequency of the series resonant tank. This means the current could reach very high level during over load situation. This characteristic makes over load protection design for LLC resonant converter very critical. During over load condition, the load of the converter increases. The worst scenario will be short circuit of output. In this part, the impact of short circuit output will be investigated for LLC resonant converter. The simulation waveforms of LLC resonant converter during normal operation and over load operation are shown in Figure 5.24. From the simulation waveforms, lost of ZVS and high current stress could be observed during over load condition. This could be understood through the characteristic of the

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converter. In Figure 5.25, the DC characteristic of LLC resonant converter is shown. At normal operation, the converter is working at Point A, when over load condition happens, the operating point will move to Point B. As seen in the graph, point A is in ZVS region while point B is in ZCS region. The over load current for different switching frequency is shown in Figure 5.26, the over load current could rise to very high.

(a)

(b) Figure 5.24 Simulation waveforms of LLC resonant converter at (a) normal operation, and (b) short circuit operation

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Figure 5.25 Lost of ZVS for LLC resonant converter during over load situation

Figure 5.26 High current stress during over load situation for LLC resonant converter

From these results, the major problems for LLC resonant converter during over load condition are: high current stress, and lost of ZVS.

5.2.2 Method 1: Increasing Switching Frequency

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When converter running into over load protection condition, there are two ways to limit the current. First way is to reduce the average voltage applied to the converter. For example, in PWM converter, duty cycle is reduced to limit the current. By reducing duty cycle, the average voltage applied to converter is reduced so that the current can be limited. Second way is to increase the impedance of the power stage of the converter so to limit the current. This method is useful for variable frequency controlled converters. By moving the switching frequency away from resonant frequency, the impedance of the resonant tank will increase so that the current can be limited. To simplify the problem, let’s look at the worst scenario: short circuit of output. Under such condition, the LLC resonant converter could be simplified into a simple series resonant tank as shown in Figure 5.27.

Figure 5.27 Simplified model of LLC resonant converter during short circuit condition

With this model, the switching frequency needed to limit the output current during short circuit situation could be derived. It is shown in Figure 5.28. As seen

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in the graph, if the desired over load current is 27A, then the switching frequency need to increase to about 400kHz.

Figure 5.28 Short circuit output current at different switching frequency

Figure 5.29 shows the average for different over load condition. From Fig.2 we can see, by moving switching frequency away from resonant frequency (250kHz), the output current can be limited. There are two directions to move switching frequency: move to higher frequency or lower frequency in relationship to resonant frequency. Since the lower frequency will result in ZCS condition as shown in Figure 5.30, which is not a desirable working condition for MOSFET, here we will move switching frequency to higher frequency.

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Figure 5.29 Average output current vs. switching frequency under short circuit

Figure 5.30 Change of operating mode with different switching frequency under protection mode

Figure 5.31 shows the test waveforms for this condition. In the real test, because of the parasitic parameters, with 358kHz switching frequency, the output current can be limited under 27A under short circuit condition.

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Figure 5.31 Test waveform (top to bottom: Q1 gate signal, Transformer primary current and resonant cap voltage)

Figure 5.32 Problems with high switching frequency protection mode

For this method, the converter will be working at pretty high switching frequency during over load protection mode compare with normal operation condition. With high switching frequency, there are several considerations: First the switching loss will increase. As shown in Figure 5.32, during short circuit condition, current stress reaches the highest. Turn off current also reaches

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the highest. With so high switching frequency, the loss on the device will be very high which will increase the thermal management requirement. Second, the stress on the magnetic components will be very unbalanced. During over load protection, switching frequency reaches highest level while all the volt-second is applied to the inductor, which means inductor has to be designed according to this highest. For LLC resonant converter, this frequency will be almost double of normal operation frequency; this will make the size of the inductor to be larger.

5.2.3 Method 2: Variable frequency control plus PWM Control From previous discussion, reduce the voltage applied to the converter can limit the current too. In the second method, variable frequency control and PWM control method are combined. For this method, the converter has two modes: normal operation mode and protection mode. During normal operation mode, variable frequency control is used to get high efficiency. During over load protection mode, first switching frequency is increased to limit the current, when switching frequency reaches the limit we set, PWM control mode will be used to reduce the voltage applied to resonant tank as shown in Figure 5.33. With this method, the output current can be effectively limited. As shown in Figure 5.34, the current can be limited with duty cycle change. In this graph, when switching frequency is lower than 300kHz,

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variable frequency control is used. When switching frequency reaches 300kHz, duty cycle control cut in.

Figure 5.33 Control Method of Variable freq + PWM control

Figure 5.34 Average output current of LLC converter with variable frequency + PWM control

In Figure 5.34, a flat area is observed when duty cycle is close to 0.5. In this flat area, the duty cycle change cannot change the current. The reason is for each switching cycle, the body diode of the MOSFET will conduct for some time; duty

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cycle change must be larger than the body diode conduction time as shown in Figure 5.35.

Figure 5.35 Simulation waveform for D=0.5, 0.4 and 0.2 at short circuit condition

Figure 5.36 shows the simulation result with consideration of output capacitance of MOSFET. The current will resonant instead of stay at zero. Also can be seen from Fig.10 that the ZVS condition of MOSFET is lost because of DCM operation of primary current. Figure 5.37 shows the test result.

Figure 5.36 Simulation waveform with PWM control (from top: gate signal of Q1 and Q3, Vds of Q1 and primary current Ip)

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Figure 5.37 Test waveform for PWM control (Top: Vds of Q1, middle: gate signal of Q1, and Q2, bottom: primary current)

This method can achieve the current limiting function. The concern is operating condition. Since ZVS is lost during over load protection mode, the switching loss will increase and noise on gate driver will be a problem too. Another issue will be how fast the transition between different modes could be. Since the current could ramp up very fast, a very fast protection is necessary.

5.2.4 Method 3: LLC resonant converter with clamping diode In this method, the current limit function is built in the power stage. This method can provide cycle-by-cycle current limiting function without control interference. Also, this method provides some other benefits too. Next the detail of this method will be discussed. Figure 5.38 shows the original LLC resonant converter and proposed LLC resonant converter. The proposed LLC resonant converter is different from previous discussed LLC resonant converter in following aspects: first, the resonant capacitor is spited into two capacitors. Then, two diodes are put in 173

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parallel with the resonant capacitor. With this modifications, there are several benefits could be achieved.

(a)

(b) Figure 5.38 Two LLC resonant converter topologies: (a) Original LLC converter and (b) proposed clamped LLC converter

First benefit is achieved through splitting the resonant capacitor. Figure 5.39 shows the simulation waveforms of these two topologies. As seen from the simulation waveform, with splitting resonant capacitor, the input current will have lower ripple. This will alleviate the stress put on the high voltage bus capacitor.

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(a)

(b) Figure 5.39 Simulation waveforms for two LLC resonant converter topologies: (a) original LLC converter and (b) clamped LLC converter

Another benefit will be over load protection, which is provided by the clamping diodes. Figure 5.40 shows the simulation waveforms of original LLC resonant converter and the clamped LLC resonant converter at over load condition. For original LLC resonant converter, it can be seen that during over load condition, input current is very high and the peak voltage across resonant capacitor will increase to very high too. This is because during over load condition, more current is going through the resonant tank, which will charge the resonant cap to higher voltage. For LLC resonant converter with clamping diodes,

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first the voltage stress on resonant cap is limited so that a low voltage cap can be used; another benefit is that by limit the voltage on resonant cap, the energy could be absorbed by resonant tank is limited as shown in the state plane in Figure 5.41. Also could be observed from the simulation waveform of clamped LLC resonant converter, under clamped operating mode, ZVS is still achieved.

Figure 5.40 Simulation waveforms under over load condition for (a) original LLC converter and (b) clamped LLC converter

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Figure 5.41 State plane of original and clamped LLC resonant converter

The over load current for both topologies are shown in Figure 5.42 and Figure 5.43.

Figure 5.42 Average output current under over load condition for original LLC converter

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Figure 5.43 Average output current under over load condition for clamped LLC converter

Another benefit of this method is that this method doesn’t need active control; it is very simple to implement. Its response speed is fast, which can provide cycleby-cycle current protection. During normal operation, these two diodes will not conduct, the clamped LLC converter operates exactly same as original LLC resonant converter in every aspects. In order to avoid the clamp diodes to impact normal operation condition, the design is chosen as shown in Figure 5.39. Within the expected operating region of the converter, the voltage stress on resonant capacitor is designed to be lower than the clamping voltage. Figure 5.44 shows the design region for clamped LLC resonant converter. During normal operation condition, the voltage stress on the resonant capacitor is always lower than the clamp voltage, which is the input voltage. Figure 5.45 shows the test waveforms with this method. With this method, the converter is tested with short circuit with output current at 32A at switching frequency at 200kHz for over 5 minutes. 178

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Figure 5.44 Design region of clamped LLC resonant converter

Figure 5.45 Test waveform of LLC converter under clamping mode

To use this method, there are several concerns. As described before, because of these clamping diodes, the current is limited for each switching cycle. The current can be passed through the resonant tank is related to the input voltage. Also, since this method limit the amount of current flow through resonant tank in

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each switching cycle, when switching frequency is changing, the average output current will change too. Let’s look at an example next. For the given application, when input voltage is 300V, we set the maxim output current at 27A. When input voltage is 400V, two things changes: input voltage is higher, switching frequency is higher too. From above analysis, this will increase the maxim output current. Instead of 25A at 300V, the maxim output current at 400V will be 34A. Although with this drawback, the clamping diode is still an effective way to protect the converter. With these clamping diodes, ZVS is ensured at all condition. At high input voltage, although the setting point increased, still it gives us enough time to let the controller to take over and limit the current. Base on this information, the compensator could be designed and the front end DC/DC converter is a complete system now.

5.3 Integrated power electronics module for LLC From above analysis and test results, LLC resonant converter demonstrated significant improvements over PWM topologies. With high frequency and high efficiency, the power density of LLC resonant converter is improved by 3 times compared with asymmetrical half bridge. As seen in chapter 3, with advanced packaging technology, the power density and performance of asymmetrical half bridge converter could be improved 180

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significantly too. In this part, integrated power electronics module for LLC resonant converter will be discussed. For active IPEM, it is the same for both asymmetrical half bridge converter and LLC resonant converter. With smaller turn off current and loss on the active IPEM, the thermal stress on active IPEM in LLC resonant converter will be much less. This could results to reduced thermal requirement. For the passive IPEM for LLC resonant converter, it is different from asymmetrical half bridge converter. As discussed in previous part, with splitting resonant capacitor and clamping diodes as shown in Figure 5.46, current limiting and smooth input current could be achieved. From here, the passive IPEM for LLC resonant converter could be identified. The passive IPEMs for asymmetrical half bridge and LLC resonant converter are shown in Figure 5.47.

Figure 5.46 LLC resonant converter with splitting resonant cap and clamping diodes

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Figure 5.47 Schematics of passive IPEM for AHB and LLC r converter

Comparing these two passive IPEMs, they are very different in several ways. First, passive IPEM for AHB consists two transformers. In LLC passive IPEM, only one transformer with center-taped secondary is needed. Second, the series inductor and capacitor have very different value. For LLC resonant converter, the capacitor is around 40nF while AHB need 1uF capacitor. Third, for LLC passive IPEM, two capacitors are needed to utilize clamping LLC topology. For asymmetrical half bridge, only one capacitor is integrated. To integrate two resonant capacitors into the structure, another dielectric layer is used as shown in Figure 5.48

Figure 5.48 Capacitor integration for LLC resonant converter

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There are different structures to build passive IPEM for LLC resonant converter. Two structures are shown in Figure 5.49.

Figure 5.49 Two passive IPEM structures for LLC resonatn covnerter

For the fist method, it uses one planar E core and an I-core. It is build with transformer with controlled leakage by inserting a leakage layer between primary winding and secondary winding. The resonant capacitor could be constructed by building primary winding on dielectric material. The other method use similar structure as asymmetrical half bridge. With two planar E core and one I core, with integrated magnetic concept, all the magnetic components could be integrated into this structure. The resonant inductor winding will be built on dielectric layer to provide resonant capacitors. Comparing these two methods, first method is simpler. The issues of this structure are: first, accurately control the leakage inductance is not easy. For resonant converter, resonant inductance value to the operating point of the converter. The value of the inductance needs to be accurately controlled. With this method, the leakage inductance could not be very accurately controlled. Second, with this structure, interleaving of winding is impossible. For high frequency

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operating, this could introduce high winding loss in the structure. With method two, these problems could be solved with more complex structure. With advanced packaging technology, all the passive components except output filter capacitor could be integrated into the planar structure. Also, the active IPEM will reduce the size for primary switches. With advanced integration, the power density of 400kHz LLC resonant could be further improved.

Figure 5.50 Power density of discrete LLC and projected integrated LLC

For PWM converter, the high thermal stress and requirement of snubber prevented from integrate secondary rectifier diodes into passive IPEM. With LLC resonant converter, first the thermal stress on secondary rectifier diodes is greatly reduced; also, with shottoky diodes and natural commutation, there is no need for snubber. This gives us opportunity to integrate the secondary switches into passive IPEM. With this integration, first the system will be built with just two blocks, which makes the system very simple. With this method, the parasitic between rectifier and transformer could be minimized which will be beneficial for high switching frequency operation.

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5.4 Summary In this part, the over load protection issue of LLC resonant converter is been investigated. Three ways of over load protection methods are discussed. For each method, there are some benefits and concerns. Increasing switching frequency is simple to use and implement. The main concern is that magnetic design will be greatly affect by how high the frequency will be. Also, during protection, current stress is very high for primary switches. The thermal design for primary switch will be suffered to deal with this condition. For variable frequency + PWM control, some modification on the control circuit is necessary to implement it. This method will prevent the issues of high frequency operation in method a. We can choose a lower frequency and use PWM control to limit the current so that magnetic and semiconductor doesn’t to be over designed. The problem of this method is that during protection mode, primary switches will loss ZVS. LLC resonant converter with splitting cap and clamping diodes is a very effective way to limit the output current during over load condition. Basically this is a passive method to limit the current. With splitting cap, input current ripple can be reduced greatly. With clamping diodes, the current at over load condition

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can be automatically limited. The voltage stress on the resonant cap is also kept under a given voltage. ZVS is achieved during over load protection mode. The problem of this method is that the setting point is a function of input, output voltage. So for different operating point, this setting value will change. It reaches minimal at low line and high output voltage and reaches maximum with high line and low output voltage. Since each of these three methods has its pros and cons, for different requirement, different over load protection method should be choose. In some case, combination of different method could be used to get better performance too.

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Chapter 6 Small signal analysis and control design of LLC converter

6.1 Introduction In previous chapters, the characteristic, design and advantages of LLC resonant converter were discussed. As demonstrated in chapter 3, LLC resonant converter has very low switching loss. Because of low voltage stress on secondary rectifier, low voltage rated diodes could be used, conduction loss is also much reduced compared with PWM converter. With DC analysis and understanding of the operation of LLC resonant converter, power stage parameters could be designed to meet given specifications. To use LLC resonant converter as front end DC/DC converter, still another important issue need to be investigated: small signal characteristic. Small signal characteristic is essential for the feedback loop design. For front end DC/DC converter, feedback control is needed to provide a tight regulation of output voltage with load and input variation, which happens all the time for front end DC/DC converter. In Figure 6.1, the whole converter with control circuit is shown. For LLC resonant converter, variable frequency control is used. To achieve variable frequency control, instead of PWM comparator in PWM controller, a Voltage Controlled Oscillator (VCO) is used to convert control 187

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voltage Vc to the variable frequency square wave, which is used to drive the switches. To design the compensator, we have to know the small signal characteristic of the converter. In this part, the small signal characteristic of LLC resonant converter with VCO will be investigated. Base on the small signal characteristic of LLC resonant converter, the compensator design will be investigated later.

Figure 6.1 LLC resonant converter with feedback control

For PWM converter, state space average method has been widely used. State space average method provides simple and accurate solution for up to half switching frequency. It has been verified and the theoretical system has been well established. With the small signal model derived from state space average method, small signal characteristic of PWM converter can be studied and control circuit can be designed accordingly.

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Unfortunately, state space averaging method cannot be applied for frequency controlled resonant converter. This is because of the totally different ways of energy processing methods for these two kinds of power converter. For PWM converter, the natural frequency of the linear network (output filter) is much lower than the switching frequency. The modulation of the converter is achieved through the low frequency content in the control signal. With this character, the average method can provide approximate linear solution of the nonlinear state equations. The derived model has a continuous form and is accurate up to half of switching frequency. However, for resonant converter, the switching frequency is close to the natural frequency of the linear network (resonant tank). The states contain mainly switching frequency harmonics instead of low frequency content in PWM converter. The modulation of the resonant converter is achieved by the interaction between switching frequency and resonant frequency. Since average method will eliminate the information of switching frequency, it cannot predict the dynamic performance of resonant converter [D-6][D-7]. In the past, several methods were tried to solve this problem. Among these methods, some made too many simplifications that the results cannot match with test results. Some of them are very complex and difficult to use [D-8][D-9]. In this dissertation, two methods were used. One is Extended Describing Function method developed by Dr. Eric X. Yang. This method is a simplified version of describing function method. A software package in Matlab is also

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developed to realize this method. With the software package, small signal characteristic of a converter could be derived with short simulation time. Another method used in this dissertation is a simulation-based method. This method uses simulation tools to emulate the function of impedance analyzer to get the small signal response of the converter. The method is based on time domain switching model simulation, which is a necessary for every converter design. So no extra modeling effort is needed for this method. It could be used to any periodical operating converter. It is a very effective method to deal with complex topology, which is difficult to deal with conventional method. Also, the impact of parasitic could also be easily included into this method. This chapter is organized in following way. First, two methods: extended describing function method and simulation-based method, will be introduced. With these two methods, small signal characteristic of LLC resonant converter will be studied. Load impact, and resonant tank value impact will be studied with these tools. Finally, the results from these two methods will be compared with test results. With the information of small signal characteristic of LLC resonant converter, the design of the compensator will be discussed.

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6.2 Extended Describing Function analysis Dr. Eric X. Yang published extended describing function method in [D-12]. This method is a simplified modeling method based on the describing function method published by J. O. Groves [D-9]. With this method, the small signal model of a periodical operating converter could be derived with any order of harmonics of switching frequency taken into consideration. This method could be used for PWM converter. With only DC components of state variables taken into consideration, it is same as state space averaging method. For resonant converter, since switching frequency and its harmonics also play important roles in the power transfer process. State space averaging method could not be applied. With extended describing function, high order harmonics could be included so that an accurate model could be derived. The detail of extended describing function method and introduction of the software package could be found in [D-12]. The process of building the model for extended describing function is discussed in Appendix D. The model file of LLC resonant converter needed to perform the analysis are attached in Appendix D too. In next part, the small signal characteristic of LLC resonant converter will be discussed using extended describing function method. The circuit parameters used for this analysis is shown in Figure 6.2.

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Figure 6.2 Circuit parameters for extended describing function analysis

For extended describing function method, the order of harmonics needed for accurate model is one thing needs to be determined before doing the analysis. For traditional resonant topologies like SRC and PRC, only the fundamental harmonic of switching frequency will be sufficient to provide an accurate small signal model [D-11][D-12]. For LLC resonant converter, though, it is a multi resonant converter. The fundamental component of switching frequency might not be enough. In Figure 6.3, the control to output transfer function is shown for region 1 (switching frequency higher than series resonant frequency). As seen from the graph, in region 1, fundamental component seems to be enough. With higher order of harmonics took into consideration, the model will not be improved significantly. This is understandable since in region 1, LLC converter operates very similar to SRC.

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Figure 6.3 Impact of harmonic order on the accuracy of EDF method in region 1

In Figure 6.4, same analysis was done in region 2 (switching frequency lower than series resonant frequency). In region 2, fundamental component is not enough. With more harmonics considered, the model will be different from only consider the fundamental component. But after the 5th harmonic, include more harmonics doesn't make any significant difference anymore. In the later simulation, we will use 1st, 3rd and 5th harmonic for analysis. This result is also reasonable because in region 2, LLC resonant converter is working as a multi resonant converter. During each switching cycle, the resonant frequency changes as topology modes progress.

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Figure 6.4 Impact of harmonic order on the accuracy of EDF method in region 2

With up to 5th harmonic take into consideration; the small signal characteristic of LLC resonant converter is derived with extended describing function method. With this requirement, the simulation time is extended. Another problem with extended describing function method is that to build the model, every operating modes of the circuit need to be identified. For LLC resonant converter, it has many different operating modes as shown in Appendix B. It would be very difficult to build the model file. Next, time domain simulation-based method will be discussed, which could solve these problems.

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6.3 Time domain simulation method This method uses brute force simulation to derive the small signal characteristic of LLC resonant converter. It emulates the function of a network analyzer. To perform this analysis, only the switching model of the converter is needed, there is no other model needed, which makes this method very attractive. The procedure of this method is shown in Figure 6.5.

Figure 6.5 Procedure for simulation method to analyze small signal characteristic

First step of this method is to simulate the converter at given operating point (Load condition, switching frequency and input voltage) without perturbation as shown in Figure 6.6. After simulate to steady state, record all the information needed as the base information. Second step is to simulate the converter with perturbation added to where interested. For example, to investigate the control to output characteristic, a perturbation will be added to the control voltage as shown in Figure 6.7. This

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perturbation will be a small amplitude sinusoidal signal with known phase information. The amplitude is small so that the converter operation modes will not change with perturbation added. With perturbation injected, make another time domain simulation to steady state and record all the information interested.

Figure 6.6 Circuit setup for first step simulation

Figure 6.7 Circuit setup for second step simulation

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Next, the results of previous two simulations will be compared. The impact of the injected perturbation on output variable could be derived. This will give us the small signal characteristic of the converter at one perturbation frequency. Repeat above steps for the frequency range interested, a complete small signal characteristic at given operating condition could be extracted. If other operating point is interested, change the switching circuit model so that the converter is operated at new operating point. As can be seen, this method asks for extensive simulation power. Fortunately, with advanced software and computer, this is not so time consuming a method. First, with Simplis software, above process could be automated. The software could do the sweeping of frequency and operating condition as set. It also performs the extraction of small signal characteristic after each simulation. With this software, one bode plot of the converter at given operating condition could be simulated in two hours. With simulation method, a SRC was analyzed. The results were shown in Appendix C.

6.3.1 Small signal characteristic of LLC resonant converter With extended describing function method, the characteristic in region 1 is shown in Figure 6.8. It is a three poles and one zero system.

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As seen from the graph, in region 1, there are one beat frequency double pole, one low frequency pole and one ESR zero. As switching frequency moves close to resonant frequency, the beat frequency double pole will move to lower frequency. When the switching frequency is very close to resonant frequency, the beat frequency double pole will eventually split and becomes two real poles. One moves to higher frequency and one move to lower frequency as switching frequency continuous move close to resonant frequency. Finally, the pole moves to low frequency will combine with the low frequency pole caused by the output filter and form a double pole. This characteristic is same as could be observed in SRC converter. In this analysis, the ESR of output capacitor is considered. This ESR will introduce an ESR zero at fixed frequency.

Figure 6.8 System poles and zeros of LLC in region 1 with different switching frequency 198

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Audio susceptibility, input conductance and output impedance in region 1 are also shown in Figure 6.9, Figure 6.10, and Figure 6.11.

Figure 6.9 Input conductance of LLC converter in region 1

Figure 6.10 Output impedance of LLC resonant converter in region 1

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Figure 6.11 Audio susceptibility of LLC converter in region 1

The characteristic in region 2 is shown in Figure 6.12. In this region, the system has some very different characteristic. A Right Half Plane Zero is observable in this region. This RHZ moves with switching frequency. Fortunately, this RHZ doesn’t shift to very low frequency region even when switching frequency is very low. This is good since it is not easy to deal with the RHZ. In left half plane, there are three poles and one zero. They are pretty stable compared with poles and zero in region 1. In region 1, when switching frequency moves close to resonant frequency, one pole moves to higher frequency. When the converter runs into region 2, as switching frequency further reduces, this pole will move back to lower frequency, but not so much. In this region, the switching frequency has less impact on the double pole at low frequency and no impact on the ESR zero. 200

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Figure 6.12 System poles and zeros of LLC converter in region 2

Audio susceptibility, input conductance and output impedance in region 2 are also shown in Figure 6.13, Figure 6.14, and Figure 6.15.

Figure 6.13 Input conductance of LLC resonant converter in region 2

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Figure 6.14 Output impedance of LLC resonant converter in region 2

Figure 6.15 Audio susceptibility of LLC resonant converter in region 2

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From above analysis results, the small signal model of LLC resonant converter could be extracted. In region 1, this converter is very similar to the series resonant converter. In region 2, though, it is very different. One RHZ could be observed in region 2. The poles and zero in left half plane are very stable with the changing of switching frequency, which is very different from normal resonant converter. The problem of this method is that to get accurate small signal model of the converter, a good model file is needed. This is a very time consuming process especially when the converter could run into many different operating modes. Another problem is that the accuracy is depends on the order of harmonics took into consideration. With higher order of harmonics, the simulation time and convergence problem will be difficult to deal with. Due to the difficulties to build the model file, it is not easy to take the parasitic components into consideration. In next part, the time domain simulation method will be discussed. Next, simulation based method will be used. The simulation is performed on LLC resonant converter as shown in Figure 6.16. The resonant frequency of Cr and Lr is designed at 250kHz. Here full load condition is used to analyze the small signal characteristic. Later load impact will be investigated.

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Figure 6.16 LLC converter setup for small signal analysis

In Figure 6.18, the small signal characteristic of LLC resonant converter is shown. The simulation is performed for a switching frequency range from 100kHz to 400kHz to cover all three operating regions. In the small signal characteristic of LLC resonant converter, three distinctive regions exist correspond to the three operating regions shown in the DC characteristic. Next the characteristic of these three regions will be discussed in detail. For region 1, the converter operates similar as a series resonant converter. The small signal characteristic is also very similar to SRC. Low frequency pole and beat frequency double pole could be observed in this region.

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Figure 6.17 Operating region of LLC resonant converter

Figure 6.18 Bode plot of control to output transfer function for LLC resonant converter

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Figure 6.19 Bode plot of control to output transfer function of LLC resonant converter in region 1

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Figure 6.20 Bode plot of control to output transfer function of LLC resonant converter in region 2

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The characteristic in region 2 is shown in Figure 6.20. Region 2 is a very interesting region. In this region, the DC characteristic is like a PRC. But for the small signal characteristic of LLC resonant converter is very stable in this region. As seen in the graph, there is no beat frequency double pole. As switching frequency changes, the characteristic doesn't change much. At low frequency, instead of single pole, now it is a double pole. This double pole moves as switching frequency changes. Since the switching frequency range is not so wide, with in region 2, this double pole doesn't move too much. There is a sign of a right half plane zero exists in this region though. From the graph, it can be seen that at 30k to 40kHz frequency range, the magnitude of the characteristic changes slope from -40dB/Dec to -20dB/Dec while the phase is continue reducing. For front-end application, the bandwidth is normally designed at 2 to 5kHz. This right half plane zero shouldn't impact too much on the feedback loop design. Region 3 is ZCS region, which is not a desired operating region for this application. From the simulation results, following observation could be made: 1. There is no beat frequency dynamic problem at the boundary between region 1 and region 2. This gives us opportunity to operate the converter right at

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the resonant frequency of Cr and Lr, which is boundary point between region 1 and region 2. 2. In region 1, the converter behaves very similar to SRC. Beat frequency double pole and low frequency pole could be observed. 3. In region 2, the small signal characteristic of the converter is pretty stable with switching frequency change. 4. Between region 2 and region 3, beat frequency dynamic could be observed. The phase of small signal characteristic will jump for 180 degree across the boundary. Above analysis is performed at given load. Next the impact of load change on the small signal characteristic will be investigated.

6.3.2

Impact of load variation on small signal characteristic

In this part, the impact of load variation on the small signal characteristic of LLC resonant converter will be investigated. The simulations were performed in region 1 and region 2. The small signal characteristic of LLC resonant converter with different load in region 1 (fs=300kHz > fr=250kHz) is shown in Figure 6.21.

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Figure 6.21 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 1(fr=250kHz, fs=300kHz)

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Figure 6.22 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 1(fr=250kHz, fs=300kHz) (full load to 25% load)

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Figure 6.23 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 1(fr=250kHz, fs=300kHz) (25% to no load)

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From the graph, several things could be observed. With load changes, the small signal characteristic of LLC resonant converter could be divided into two regions as shown in Figure 6.22 and Figure 6.23. In the first region, the characteristic doesn't change much. Within the region, the converter still works in continuous conduction mode. When load reduced to some level, the converter will run into DCM as discussed in Appendix B. Then the low frequency pole will move to lower frequency and beat frequency double pole will move to higher frequency. At light load, LLC resonant converter could be treated as a first order system in very wide frequency range. The small signal characteristic of LLC resonant converter with different load in region 2 is shown in Figure 6.24. It could be divided into three load ranges according to different trends in the moving direction of poles and zeros as shown in Figure 6.25, Figure 6.26 and Figure 6.27. In first load range, as load decreases, the Q of low frequency double pole will reduce. The right half plane zero tends to move to higher frequency and eventually move out of half switching frequency range. In the second load range, however, the quality factor of low frequency double pole will increase as load further decrease. As load continue reduce, the characteristic will come into load range 3. In load range 3, the low frequency double pole will split. One move to low

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frequency and one move to high frequency, just as could be observed in PWM converter.

Figure 6.24 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz)

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Figure 6.25 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) (full load to 25% load)

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Figure 6.26 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) (25% to 10% load)

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Figure 6.27 Bode plot of control to output transfer function of LLC resonant converter with load variation in region 2(fr=250kHz, fs=200kHz) (10% to no load)

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From above simulation results, one conclusion is that with light load, one low frequency pole will exist. This needs to be considered when design the compensator.

6.4 Impact of circuit parameters In this part, the impact of some components value on the small signal characteristic of LLC resonant converter will be shown. The components will be investigated include: output filter capacitor, magnetizing inductance Lm, and resonant tank impedance.

6.4.1 Impact of output capacitance In this part, the small signal characteristic of LLC resonant converter with different Co will be simulated.

Figure 6.28 Simulation setup for output capacitor impact on small signal characteristic

The converter is shown in Figure 6.28, the resonant frequency is 250kHz. The simulation will be performed in two switching frequency. One frequency is in region 1 at 300kHz as shown in Figure 6.29. The other simulation is performed in region 2, with switching frequency at 200kHz as shown in Figure 6.30. 218

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From both simulation, Co only impact the low frequency pole and doesn’t affect high frequency poles.

Figure 6.29 Bode plot of control to output transfer function with different output capacitance with switching frequency 300kHz(region 1)

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Figure 6.30 Bode plot of control to output transfer function with different output capacitance with switching frequency 200kHz(region 2)

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6.4.2

Chapter 6. Small signal analysis of LLC resonant converter

Impact of magnetizing inductance

In this part, the small signal characteristic of LLC resonant converter with different Lm will be simulated. The converter been simulated is shown in Figure 6.31, the resonant frequency is 250kHz. Same as for previous case, two switching frequency points will be choose. One frequency is in region 1 at 300kHz as shown in Figure 6.32. The other simulation is performed in region 2, with switching frequency at 200kHz as shown in Figure 6.33.

Figure 6.31 Simulation setup for magnetizing inductance impact on small signal characteristic

From the simulation in region 1, Lm doesn’t affect the small signal characteristic in this region at all. With Lm changed by 10 times, the small signal characteristic is almost constant. In region 2, Lm has great impact on the DC gain of the small signal characteristic. With larger Lm, the right half plane zero also tends to shift to lower frequency.

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Figure 6.32 Bode plot of control to output transfer function with different magnetizing inductance with switching frequency 300kHz(region 1)

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Figure 6.33 Bode plot of control to output transfer function with different magnetizing inductance with switching frequency 200kHz(region 2)

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6.4.3

Chapter 6. Small signal analysis of LLC resonant converter

Impact of resonant tank impedance

In this part, the small signal characteristic of LLC resonant converter with different resonant tank impedance will be simulated. The resonant frequency is kept constant in the simulation. The converter been simulated is shown in Figure 6.34, the resonant frequency is kept constant at 250kHz, which means as Lr been changed, Cr will be changed accordingly. Same as for previous case, two switching frequency points will be choose. The 300kHz case is shown in Figure 6.35. The 200kHz case is shown in Figure 6.36.

Figure 6.34 Simulation setup for resonant tank impedance impact on small signal characteristic

As from the simulation, in region 1, as impedance of resonant tank increases, which means increase Lr and reduce Cr, the DC gain will increase. This is understandable since with higher impedance, the Q with given load will increase, then the slope of the DC characteristic will have larger value, which is the DC gain in small signal characteristic. Another interesting thing is that the first pole will move with different resonant tank impedance, which means in LLC resonant converter, the lowest pole is not determined by output filter only. In region 2, the

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similar impact on DC gain could be observed. With larger Lr, one low frequency pole also moves to higher frequency.

Figure 6.35 Bode plot of control to output transfer function with different resonant inductance with switching frequency 300kHz(region 1) 225

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Figure 6.36 Bode plot of control to output transfer function with different resonant inductance with switching frequency 200kHz(region 2)

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6.5 Test verification In this part, a test circuit was built with same parameters as used in the analysis. The test setup is shown in Figure 6.37.

Figure 6.37 Test setup up for small signal characterization of LLC converter

In Figure 6.38, the results in region 1 with full load are shown for three methods: test, simulation and extended describing function. From the comparison, these three results match each other very good. In Figure 6.39, the results in region 2 with full load are shown for three methods: test, simulation and extended describing function. From the comparison, these three results match each other very good. From the verifications, both methods match test results very well. These two methods have their pros and cons. For simulation method, it is easy to implement. With powerful computer and software, it is also fast. The problem is lacking of 227

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insight of the model of the converter. It just gives the bode plot of the characteristic of the converter. If more information is needed, extended describing function method could be helpful. With extended describing function method, more information about the small signal characteristic of the converter could be derived. The drawback is that to build the model, a thorough understanding of the converter is critical. When the operating modes of the converter are too complex, this will be a painful process.

Figure 6.38 Bode plot of control to output transfer function at full load in region 1

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Figure 6.39 Bode plot of control to output transfer function at full load in region 2

6.6 Compensator design for LLC resonant converter From above analysis, we have a complete picture of the small signal model of LLC resonant converter. Base on this information, the compensator could be designed. First, as seen in the characteristic of LLC resonant converter, the phase at DC is 180-degree instead of 0-degree as seen for PWM converter. This means from the control voltage point of view, LLC resonant converter is an inverter. As 229

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control voltage increases, output voltage will decrease. This is because of the fact that for resonant converter to work under ZVS condition, the output voltage will decrease when switching frequency increases. For voltage-controlled oscillator, when its input voltage increases, the frequency will increase. For PWM converter, duty cycle will increase as control voltage increases, which will increase the output voltage. For PWM converter, the compensator is a negative feedback as shown in Figure 6.40. For LLC resonant converter, a positive input compensator is needed as shown in Figure 6.41 because of the negative transfer function of the converter.

Vc1 = −

Z2 Vo1 Z1

Figure 6.40 Compensator for PWM converter

Vc 2 =

Z2 Vo 2 Z1

Figure 6.41 Compensator structures for LLC resonant converter

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For LLC resonant converter, its designed operating region is region 2 (switching frequency lower than series resonant frequency). In this region, the small signal characteristic of LLC resonant converter is pretty stable with changing of switching frequency. Although a RHZ exists in this region, it never moves to very low frequency. The more significant impact is the load change. With light load, one pole will move to very low frequency. With integrator in the compensator, this might introduce conditional stable situation.

Figure 6.42 Small signal characteristic of LLC converter in region 2

Figure 6.43 Load impact on small signal characteristic of LLC converter in region 2

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Although region 2 is the designed operating region, converter might operate in region 1 due to the fact that the intermediate bus is loosely regulated. Load or AC line transient could cause this voltage rise to as high as 430V. During those conditions, the converter will operate in region 1. So the characteristic in region 1 also needs to be considered during compensator design. In region 1, the converter will have a beat frequency double pole and one low frequency pole.

Figure 6.44 Small signal characteristic of LLC converter in region 1

As load changes in region 1, similar phenomenon could be observed as in region 2. The double pole will split and one moves to high frequency, one moves to very low frequency.

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Figure 6.45 Load impact on small signal characteristic of LLC converter in region 1

With above information, the compensator could be designed. Since the RHZ is at pretty high frequency, it will not impact the compensator design so much. What need to be dealt with are one double pole and one ESR zero. At light load condition, as one pole will move to low frequency, the low frequency pole need to be considered. To compensate this system, a compensator with one integrator, 2 poles and 2 zeros is used. The two zeros are placed to compensate the double pole exists in the system. Another consideration is the low frequency pole due to light load. With these two considerations, one zero is placed at low frequency to prevent conditional stable from happening. Another zero is placed around the double pole. The poles are placed to compensate the ESR zero and provide more attenuation at switching frequency. The compensator is shown in Figure 6.46. With this compensator, the loop gain in different operating regions is shown in Figure 6.47 and Figure 6.48. The test results of LLC resonant converter under load change are shown in Figure 6.49 and Figure 6.50. The output voltage is within 5% regulation window during full range load step.

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Figure 6.46 Compensator designed for LLC resonant converter

Figure 6.47 Plant bode plot and loop gain bode plot in region 1

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Figure 6.48 Plant bode plot and loop gain bode plot in region 2

Figure 6.49 Test result of load change from no load to full load

Figure 6.50 Test result of load change from full load to no load

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6.7 Summary In this chapter, small signal characteristic of LLC resonant converter is been investigated. Two methods were used to perform the analysis: simulation and extended describing function method. With simulation, the small signal characteristic of the converter could be covered with any operation mode. The drawback is lack of insight of the characteristic. With extended describing function method, more information could be obtained. The drawback is the needs of develop the model file, which is not an easy task to cover all operating points. The best way is to combine the power of these two methods. Then a more accurate, more efficient and more comprehensive characteristic of any converter could be obtained. The results of these two methods match very well. They were also been verified with test setup. Base on this information, the compensator could be designed and the front end DC/DC converter is a complete system now.

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Chapter 7 Summary and Future Work The fast advance in VLSI technology made smaller, more powerful systems available. At the same time, it calls for high power density, and low profile power system. To achieve these goals, high switching frequency, high efficiency, and advanced packaging technology are the paths. It is the goal of this work to develop technology to meet and exceed these challenges.

7.1 Summary PWM topologies like asymmetrical half bridge and phase shift full bridge have been widely used for front end DC/DC application. For these topologies, two aspects limited the power density and efficiency. One is high switching loss related to high turn off current. The other one is the problem with hold up time requirement. With hold up time requirement, the input range of front end DC/DC converter is very wide. Performance at high input voltage is essential for the power density of the system since the thermal management is designed according to high input voltage performance. For PWM converter, with wide input range, the duty cycle at high input voltage will be minimal. This will cause many problems: high conduction loss, high voltage stress on semiconductor devices,

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and high switching loss. With wide input range, the performance at high input voltage is severely impacted.

7.1.1 Improvement of the state of the art topologies The primary target is to deal with hold up time problem. Two methods were discussed to solve this problem: range winding solution and asymmetrical winding solution. For range winding solution, extra components and windings are needed. With range winding, converter could be optimized for a much narrower input range so that the performance at high input voltage could be optimized. During hold up time, range winding will provide enough gain to work with low input voltage. This method could be applied to any isolated PWM topologies with similar issue. For asymmetrical half bridge, asymmetrical winding solution is a simple and effective solution. With asymmetrical winding, the DC characteristic of asymmetrical half bridge could be modified. The duty cycle at high input voltage could be extended. With extended duty cycle, lower voltage rated devices could be used. This will provide significant improvement on the efficiency. This method could only be applied to asymmetrical half bridge. Secondary conduction loss is the biggest part in total system loss, which is caused by the high forward voltage drop of rectifier. Synchronous rectifier has been widely used in low voltage high current application because of this problem.

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For front end DC/DC application, it is not so straightforward. Because of high voltage stress on the rectifier, high voltage devices are needed. When voltage rating is higher than 200V, it is difficult to find a MOSFET could outperform diode rectifier. To use 200V devices, symmetrical half bridge is chosen. Another obstacle is the reverse recovery of the body diodes of synchronous rectifier. For 200V MOSFET, the body diodes are pretty slow. It will introduce high current spike and ringing which is both dangerous and noisy. Quasi-square-wave synchronous rectification is introduced to solve this problem. With QSW operation mode, the conduction of body diodes is totally prevented. It also helps symmetrical half bridge to achieve ZVS for primary switches. This concept could be used to other application with body diode reverse recovery problem too.

7.1.2 LLC resonant converter Although above solutions could improve the efficiency of PWM converter. None of them deals with high turn off loss of the switches, which limit the switching frequency, as the result, limit the power density achievable for PWM topologies. Resonant topologies are well known for its low switching loss. Three popular resonant topologies, SRC, PRC and SPRC, were investigated for front end DC/DC application. Unfortunately, all these topologies see big penalties when design for hold up requirement. There is no significant improve of performance compared with PWM topologies.

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Although negative outcome of above analysis, the desired characteristic for front end DC/DC application could be extracted, which enable the converter working at resonant frequency at high input voltage. Three components resonant tanks were searched thoroughly. Among 15 three components resonant tanks suitable for voltage source input, resonant tank with two inductors and one capacitor seems could provide the desired characteristic. With the resonant tank, LLC resonant converter could be constructed. It is not a new topology. But the unique operating mode of this topology enable it to be optimized at high input voltage while is able to cover wide input range at the same time. Compare with PWM topologies, LLC resonant converter could reduce 40% loss at same switching frequency.

7.1.3 DC analysis of LLC resonant converter To design LLC resonant converter, DC analysis is performed. Two methods, fundamental component simplification and simulation, were used and compared. Fundamental component simplification method is simple to use and could provide the intuitive results. The problem is that when switching frequency is away from resonant frequency the error will be significant, which will prevent us to achieve optimized design. This calls for simulation method, which could provide very accurate characteristic, but with longer time to perform. To achieve optimized design, simulation method is recommended.

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7.1.4 Magnetic integration and over load protection To achieve high power density, magnetic design is very critical. Integrated magnetic can reduce the volume of magnetic components with proper design. The common problem of integrated magnetic structure is not easy to manufacture. A novel integrated magnetic structure for LLC resonant converter is proposed. With this structure, all the magnetic components of LLC resonant converter could be integrated into one magnetic structure. There is no special gapping needed, which make the structure easy to manufacture and mechanically stable. Flux ripple cancellation is also achieved with proposed integrated magnetic structure. Compare with discrete design, 30% reduction on footprint could be expected. Over load condition is a common fault condition for power converter. To make practical use of a topology, it has to be able to deal with over load situation. For LLC resonant converter, it is working close to resonant frequency; the impedance of resonant tank is very small. This makes over load protection very important function. Three methods could be used to provide over load protection for LLC resonant converter: increasing switching frequency, variable frequency plus PWM control and clamped LLC converter. First two methods is through active control the switches to change the input voltage or the impedance of resonant tank to limit the current during over load condition. The clamped LLC converter uses clamping diodes on resonant capacitor. With clamping diodes, energy could be passed through resonant tank is limited, which will limit the stress during over load condition. 241

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Chapter 7. Summary and future work

Small signal analysis of LLC resonant converter

With above discussion, the power stage of LLC resonant converter could be designed. To complete the system, feedback control is needed to provide a regulated output. Small signal characteristic of LLC resonant converter is needed to perform this task. For resonant converter, state space averaging method is no longer valid. Two methods were used to reveal this mystic: time domain simulation method and extended describing function method. Time domain simulation method is based on switching circuit model. The method tries to emulate the function of network analyzer. This method made no simplification. As long as the switching circuit model is accurate, the result will be very close to the real circuit. Extended describing function method is a mathematical method based on describing function method for nonlinear system. With describing function method, more detailed information about the system could be revealed. This method could take any harmonic components of switching frequency into consideration. With these two methods, small signal characteristic of LLC resonant converter is investigated for different switching frequency and load condition. For the small signal characteristic of LLC resonant converter, there are several interesting phenomenon. When switching frequency is higher than series resonant frequency, the converter acts very similar to SRC. Between series resonant frequency and low frequency resonant frequency, small signal characteristic is

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more stable compare with previous region. With load change, the change on small signal characteristic is similar to those could be observed in PWM converter, split of double pole could be observed. With information about small signal characteristic of the converter, compensator could be designed. With the knowledge from this work, LLC resonant converter could be designed for given specifications. From the prototypes built in CPES, LLC resonant converter shows great improvement in efficiency and power density. Compare 200kHz switching frequency design, LLC resonant converter could improve efficiency by more than 3%. Power density could be improved by almost 100%. With 400kHz design of LLC resonant converter, the power density could be improved by more than 200% compare with asymmetrical half bridge converter.

7.2 Future work 7.2.1 Passive integration for LLC resonant converter Passive components are often the limitation on volume, and cost of the system. For LLC resonant converter, with integrated magnetic technology, all magnetic components could be integrated into single magnetic structure. With planar magnetic, the resonant capacitors could also be integrated into magnetic structure. This way, all the passive components except output cap could be integrated. This integration will provide many benefits: high density, less interconnection, better electric performance.

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With passive integration, more complex resonant tank structure could be constructed. With more complex structure, other benefits could be expected as shown in Figure 7.1, Figure 7.2, and Figure 7.3. With another branch in resonant tank, the primary RMS current could be reduced.

Figure 7.1 LLC resonant tank and LLC resonant tank with passive current shaping

Figure 7.2 Simulation waveform of LLC resonant converter

Figure 7.3 Simulation waveform of LLC resonant converter with passive current shaping

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7.2.2 LLC resonant converter for other application The most significant benefit of LLC resonant converter for front end DC/DC application is that it could be optimized for high input voltage. In fact, other than this, there are several other benefits. First, the voltage stress on the secondary rectifier is minimized to two times output voltage only. Second, the output rectifier commutates naturally, there is no reverse recovery problem. Third, switching loss of LLC resonant converter could be minimized. Fourth, without output filter inductor, the transient of LLC resonant converter could be very fast. With all these advantages, LLC resonant converter is a possible candidate for other applications like isolated point of load converter too. Higher frequency operation of LLC resonant converter With LLC resonant converter, switching loss could be minimized. By control magnetizing inductance Lm, switching loss could also be controlled. This gave us opportunity to push to higher switching frequency. For some state of the art magnetic material, the optimal operating frequency could be as high as MHz. LLC resonant converter enable us to utilize these new material in front end application. The issue is how to trade off the design between magnetic loss, volume and operating region of the system.

7.2.3

Small signal modeling of resonant converter

In this work, the small signal characteristic of LLC resonant converter is been revealed. Still, a simple and easy to use model is not available yet, which is a 245

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major obstacle for people to accept and appreciate this topology. With extended describing function method, it is possible to get an equivalent small signal circuit model when only first order harmonic of switching frequency is considered. Unfortunately, third or even fifth harmonic are needed to model LLC resonant converter. There is still need to develop method to derive simple circuit model for this kind of topologies.

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Appendix A. Two and Three Components Resonant Tanks

Appendix A. Two and Three Components Resonant Tanks In this part, resonant tank with two and three resonant components will be listed and classified. Since this is a complex task, several boundaries were set to make it manageable. First, assume variable frequency control is the control method going to be used for these tanks, Second, for the resonant tank, input and output source type will be determined by tank configuration. For example, the output is must a current source for PRC since the primary side is capacitor, although PRC with voltage source could also work. This also means that there is no capacitor in parallel with input terminals since input is assumed to be voltage source type. Different input and output type is shown in Figure A.1 and Figure A.2.

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Figure A.1 Input type for DC/DC converter

Figure A.2 Input type for DC/DC converter

Third, assume the input is voltage source; output could be voltage source or current source.

A.1. Two resonant components resonant tank For resonant tank with two resonant components, there are totally eight different type of resonant tank configurations as shown in Figure A.3.

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Figure A.3 Two components resonant tanks

In this family, tank A is for series resonant converter. Tank C is for parallel resonant converter. Tank B is another form of parallel resonant converter. Tank E and Tank G requires a current source input, which is not commonly used for this application. Tank F could be used for voltage source input, but it is not meaningful since it cannot regulate power transferred to the load but increase the circulating energy. Tank H could also be applied to voltage source input, but then it is no longer a resonant topology because the resonant inductor will be clamped by input voltage source and never resonant with resonant capacitor. So for two resonant components resonant tank, tank A, B, C and D will be useful. The characteristics of these four resonant tanks are shown below. We can see that tank B has very similar characteristic with tank C that is widely used as parallel resonant converter.

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Figure A.4 DC characteristic of two components tank A

Figure A.5 DC characteristic of two components tank B

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Figure A.6 DC characteristic of two components tank C

Figure A.7 DC characteristic of two components tank D

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A.2. Three resonant components resonant tank There are many possibilities for three components resonant tank. In this part, they will be listed and classified. With three components, there are seventeen ways to connect them as shown in Figure A.8.

Figure A.8 Components configuration for three components resonant tank

With in these seventeen configurations, 15, 16 and 17 will result to a reduced order since two components could be replaced with one. For the other 14 configurations, with different resonant components, different resonant tank could be constructed. Since we are looking at three components resonant tank, the possible components used could be two Ls and one C or two Cs and one L. Three Ls or three Cs will not result to three components resonant tank and will be eliminated. With fourteen different configurations, there are 36 different resonant tanks as shown below.

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Figure A.9. Resonant tank for components configuration 1

Figure A.10. Resonant tank for components configuration 2

Figure A.11. Resonant tank for components configuration 3

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Figure A.12. Resonant tank for components configuration 4

Figure A.13. Resonant tank for components configuration 5

Figure A.14. Resonant tank for components configuration 6

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Figure A.15. Resonant tank for components configuration 7

Figure A.16. Resonant tank for components configuration 8

Figure A.17. Resonant tank for components configuration 9

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Figure A.18. Resonant tank for components configuration 10

Figure A.19. Resonant tank for components configuration 11

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Figure A.20. Resonant tank for components configuration 12

Figure A.21. Resonant tank for components configuration 13

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Figure A.22. Resonant tank for components configuration 15

These 36 resonant tanks could be classified in following table. They are classified according to the input source type and output type. For example, for resonant tank A, the input and output are directly connected, so the input and output have to be different type. In the table, it will be list as ITV or VTI, which means the input is current source and output is voltage source or vice versa. Another example is tank Y, since the input of the tank is capacitive, so input has to be current source. The output of the tank is an inductor, so output has to be voltage source. So tank Y could only be applied to ITV. For these 36 resonant tanks, there are 23 could be used for voltage source input. Next we will continue eliminate some of them. For some resonant tank here, it could not be used to regulate the output. For example, tank A could be used for VTI configuration. Since the resonant components is in parallel with the input, they cannot affect the power transferred to output, which means with this resonant tank, the output could not be regulated with variable frequency control.

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Appendix A. Two and Three Components Resonant Tanks

For some other resonant tanks, with voltage source configuration, one or more of the resonant components will not participate in controlling output power. For example, in tank I, the series resonant branch is in parallel with input. The current through this branch will not have effect on output power. So it will not behave as three components resonant converter anymore. Table A-1 Classification of three components resonant tanks Tank A B C D E F G H I J K L M N O P Q R Total

VTV

VTI V

ITV I I

ITI I

V V

I

V I V V V V

I

V V V V

I I I I I I I

I I I

Topo 1 1 2 2 3 3 4 4 5 5 6 6 7 7 8 8 9 9

VTV S T U V W X Y Z A1 B1 C1 D1 E1 F1 G1 H1 I1 J1

VTI V

V V V

ITV I I

ITI I

I V

I

V V I V

V V V 9

14

I I I I I I I 23

I I I 9

Topo 10 10 11 11 11 11 11 11 12 12 12 12 12 12 13 13 14 14

Base on above rules, tank A, E, I, J, M, A1, C1 and I1 will be eliminated. So the resonant tank could be used for voltage source input are listed as following:

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Table A-2 Three components resonant tanks with voltage source input Tank C D G H K L O S U V W X Z G1 H1

VTV V

VTI V

V V V V V V V V V V V V V 9

6

Topo 2 2 4 4 6 6 8 10 11 11 11 11 11 13 13

Next the DC characteristic of each resonant tank will be derived.

Figure A.23. DC characteristic of tank C

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Figure A.24. DC characteristic of tank D

Figure A.25. DC characteristic of tank G

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Figure A.26. DC characteristic of tank H

Figure A.27. DC characteristic of tank K

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Figure A.28. DC characteristic of tank L

Figure A.29. DC characteristic of tank O

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Figure A.30. DC characteristic of tank S

Figure A.31. DC characteristic of tank U

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Figure A.32. DC characteristic of tank V

Figure A.33. DC characteristic of tank W

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Appendix A. Two and Three Components Resonant Tanks

Figure A.34. DC characteristic of tank X

Figure A.35. DC characteristic of tank Z

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Appendix A. Two and Three Components Resonant Tanks

Figure A.36. DC characteristic of tank G1

Figure A.37. DC characteristic of tank H1

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Appendix A. Two and Three Components Resonant Tanks

In above resonant tanks, tank H, X and Z have similar characteristic as traditional called LCC resonant converter. For resonant tank G, U and W, they have the characteristic of LLC resonant converter.

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Appendix B. Operation modes of LLC resonant converter

Appendix B. Operation modes and DC analysis of LLC resonant converter In this part, different operation modes of LLC resonant converter will be discussed. LLC resonant converter, as a three resonant components resonant converter, has many different operating modes. It is a multi resonant converter. During one switching cycle, the resonant tank configuration changes. With different load condition, discontinuous conduction mode could happen. In this part, different operating modes in different operating region and load condition will be listed. From the DC characteristic of LLC resonant converter, the operating of LLC resonant converter could be divided into three regions as shown in Figure B.1. As discussed in chapter 3, region 1 and region 2 are ZVS regions, which are preferred for high frequency operation. In region 3, the converter is working under ZCS. For this converter, preferred operating regions are region 1 and region 2 in order to achieve ZVS.

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Appendix B. Operation modes of LLC resonant converter

Figure B.1 DC characteristic of LLC resonant converter

B.1. Operating modes of LLC resonant converter in Region 1 In region 1, the converter works very similar as a SRC. But because of the impact of Lm, there are some new operation modes for LLC resonant converter. In this region, there are three different operating modes as load changes.

Operating mode 1 in region 1 This mode of operation is same as a SRC. Resonant components Lr and Cr act as the series resonant tank. During whole switching cycle, Lm is clamped by output voltage and never participates in the resonant process. This mode also could be called as continuous conduction mode since the output current is always continuous. In this operation mode, Lm just acts as a passive load of series resonant tank of Lr and Cr. The operating waveforms are shown in Figure B.2.

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Appendix B. Operation modes of LLC resonant converter

Figure B.2 Waveform of operation mode 1 in region 1 for LLC resonant converter'

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Appendix B. Operation modes of LLC resonant converter

Operating mode 2 in region 1 As load becomes lighter, the converter will work into mode 2. The different of mode 2 and mode 1 is that after primary switches been switched, there will have a time period during which the secondary current is zero, or discontinuous conduction mode. During this dead time, primary current is clamped to the Lm current, the resonant tank will be consisted with Cr and Lm in series with Lr. This mode happens when following condition is met:

(VIN + VCr ) ⋅

Lm < Vo ⋅ n Lm + Lr

When above condition is met, when primary switches switched, the voltage apply to the Lm is the left of above equation. If this voltage is lower than the output voltage reflected to the primary side, the output diodes would not conduct, only after Lm current continuous charge Cr so that above condition is broken, the output filter diodes begin to conduct. Then Lm will be clamped to output voltage and no longer resonant with Cr. During this mode, Lm not only acts as the load of SRC, it also participate in the resonant with Cr. For SRC, there is no DCM in this region. The operating waveforms are shown in Figure B.3. Because of existence of Lm, there are more circulating current for LLC at light load compared with SRC. But the benefit is ZVS is ensured at light load condition.

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Appendix B. Operation modes of LLC resonant converter

Figure B.3 Waveform of operation mode 2 in region 1 for LLC resonant converter'

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Appendix B. Operation modes of LLC resonant converter

Operating mode 3 in region 1 If load continuous reduce, the converter will works into mode 3. This mode looks very similar to mode 2. But in fact, there are two kinds of discontinuous conduction modes in this operating mode. First DCM happens after the primary switches been switched as in Mode 2. But during the switching cycle, another discontinuous mode happens. From the Lr and Lm current, it can be seen that Lr current resonant and then clamped to Lm current before the switching action of primary switch. This will introduce a zero current period before primary switch action. The operating waveforms of this operating mode are shown in Figure B.4. From the waveforms, it can be seen that ZVS is still achieved because of Lm. This is the major benefit of LLC converter been discussed before. With Lm, the ZVS at light load could be maintained. Also, light load regulation is easier since Lm is always presents as the load of the SRC.

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Appendix B. Operation modes of LLC resonant converter

Figure B.4 Waveform of operation mode 3 in region 1 for LLC resonant converter'

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Appendix B. Operation modes of LLC resonant converter

B.2. Operating modes of LLC resonant converter in Region 2 This region is a very interesting region. Traditional SRC will work into ZCS when working in this region, so there is no region 2 for SRC. For LLC, with the presents of Lm, even when switching frequency is lower than resonant frequency of Lr and Cr, the converter could still work in ZVS condition with higher gain. In this region, also exist three operating modes with different load conditions.

Operating mode 1 in region 2 This is the designed operating mode for LLC resonant converter at full load. The waveforms of this operating mode are shown in Figure B.5. The major characteristic of this operating mode is the two different resonant time periods. First, when primary switches switched, Lr and Cr will resonant. During this time period, Lm is clamped by output voltage and is linearly charged. When Lr current resonant back to the same level as Lm current, second resonant happens, which is the resonant between Cr and Lm in series with Lr. This resonance will last till the primary switches been switched again. During second resonant time period, the output current keeps zero. So the output current in this operation mode is discontinuous. As seen in the waveforms, ZVS is achieved with Lm current. This gives us freedom to choose desired turn off current to achieve ZVS while with low turn off loss.

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Appendix B. Operation modes of LLC resonant converter

Figure B.5 Waveform of operation mode 1 in region 2 for LLC resonant converter'

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Appendix B. Operation modes of LLC resonant converter

Operating mode 2 in region 2 In mode 1, the output current is already discontinuous. As load becomes lighter, another discontinuous mode will exist. This discontinuous mode is very similar to operating mode 2 in region 1. When load becomes lighter, the voltage on resonant capacitor Cr will be lower when primary switches switched. If following condition is satisfied, then the secondary diodes will not conduct. This will introduce a zero current period on the output current after primary switching action.

(VIN + VCr ) ⋅

Lm < Vo ⋅ n Lm + Lr

As seen in the waveform, the energy flows back and forth from input, which means most of the current are circulating in this mode. Although this causes higher conduction loss at light load, it makes the converter operating with ZVS at very light load. This circulating current is controllable through careful design of Lm.

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Appendix B. Operation modes of LLC resonant converter

Figure B.6 Waveform of operation mode 2 in region 2 for LLC resonant converter'

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Appendix B. Operation modes of LLC resonant converter

Operating mode 3 in region 2 This is a mode in region 2 happens when the load is too heavy. In mode 1, two resonant periods exist in half switching cycle. In this mode, three modes will exist in half switching cycle. First two time intervals are the same as in mode 1. If load is too heavy, resonant capacitor Cr voltage ripple will increase. If Vcr is high enough to met following equation, the third resonant period will happen:

(VCr − Vin ) ⋅

Lm < VO ⋅ n Lm + Lr

As shown in the waveforms in Figure B.7, after Lr current resonant back to save level of Lm current, instead of clamped by Lm current, it will resonant to the other direction. This is because of too high energy is stored in resonant capacitor Cr that its voltage is high enough to make secondary diodes conduct. With this mode, the switch turn off current will be less than Lm current. The risk is that the energy is not enough for ZVS. Also, if Lr current resonant to negative, the converter will work into region 3.

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Appendix B. Operation modes of LLC resonant converter

Figure B.7 Waveform of operation mode 3 in region 2 for LLC resonant converter'

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Appendix B. Operation modes of LLC resonant converter

B.3. Operating modes of LLC resonant converter in Region 3 Region 3 is a ZCS region. In this region, the DC gain characteristic has positive slope. As seen from the waveforms in Figure B.8, switch is turned off after its body diode begins to conduct. This is not a preferred operating mode for MOSFET. In the design, this mode should be prevented.

These different operating modes will happen during the operation of LLC resonant converter. For the discontinuous operating modes, they will not affect the ZVS capability of the converter because of presents of Lm. During heavy load in region 2, the converter will come into ZCS. As discussed in over load protection, applying clamped LLC resonant topology could prevent this.

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Appendix B. Operation modes of LLC resonant converter

Figure B.8 Waveform in region 3 for LLC resonant converter

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Appendix B. Operation modes of LLC resonant converter

B.4. DC analysis of LLC resonant converter In this part, two aspects will be addressed. First is the DC characteristic. DC characteristic is the most important information for the converter design. With DC characteristic, the parameters could be chosen; the design trade offs can be made. For LLC resonant converter, the DC characteristic will draw the relationship between voltage gain and switching frequency for different load condition. Traditionally, fundamental element simplification method was used to analysis the DC characteristic of resonant converter. The fundamental element simplification method assume only the fundamental components of switching frequency is transferring energy. With this assumption, the nonlinear part of the converter like switches, Diode Bridge could be replaced with linear components. The simplified converter will be a linear network to analysis. So with this method, the DC characteristic could be derived very easily. And the result will be a close form equation, which is easy to use. For LLC resonant converter, the simplification could be done as shown in Figure B.9.

Figure B.9 Simplified topology with fundamental component assumption

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Appendix B. Operation modes of LLC resonant converter

With this simplified circuit model, the DC characteristic could be get as: Vo j ⋅ ω n ⋅ Ql = Vin j ⋅ ω (Q + 1 − 1 ) + Q (1 − ω 2 )Q ⋅ 8 n l l n s 2 2

π

ωn

ω n : Normalized switching frequency Zo :

Lr Cr

Ql : : Ratio of two resonant inductance R AC : Equivalent Load Resistance Qs:

8

π2

Lm Lr

Ro ⋅ n 2

Zo Ro

For this method, there are some limitations. Because this method is a simplified method, error will be generated with different operating point. When the current waveform is not sinusoidal and contains more high order harmonic, this method will generate high error. To evaluate this error, a more accurate DC characteristic is needed. Here simulation is used to derive the accurate DC gain characteristic. A time domain switch circuit model is built in simulation software. By changing the switching frequency and load condition, a output voltage can be get for each point. Sweep load and switching frequency, an accurate DC characteristic is got. The results of these two methods are shown in following figures. And the error is also shown.

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Appendix B. Operation modes of LLC resonant converter

Figure B.10 DC characteristic from simplified model

Figure B.11 DC characteristic from simulation method

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Appendix B. Operation modes of LLC resonant converter

Figure B.12 Error of simplified circuit model

From the error it can be seen that, when the switching frequency equals to the resonant frequency, there is no error. When the switching frequency is moving away from resonant frequency, the error will be high. This can be understood from waveform also; when the circuit works at resonant frequency, the current waveform is exactly sinusoidal, so simplified model doesn't have any error. When switching frequency is away from resonant frequency, high order harmonic content will increase, which will affect the accuracy of the simplified model. For the design of LLC resonant converter, the trade offs are more affected by operating point with maximum gain. With simplified model, large error will be introduced. Simulation gives accurate results; the issue is that it is time consuming. A better method is to combine these two methods. During primary

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Appendix B. Operation modes of LLC resonant converter

design, use simplified model to get a range. To optimize the design, simulation method is preferred to get a better design.

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Appendix C. Small signal characteristic of SRC converter

Appendix C. Small signal characteristic of SRC converter

C.1. Small signal characteristic of SRC The circuit parameters for SRC used in the simulation are shown in Figure C.1. The small signal characteristic get from simulation are shown in Figure C.2.

Figure C.1 SRC circuit for small signal analysis

In the graph, the x-axis is the frequency of the perturbation signal as in bode plot; y-axis is the magnitude in DB or phase in degree, and z-axis is the running parameter, which is the switching frequency. This is because for resonant converter, to regulate the output voltage, the switching frequency will be varied. For different switching frequency, the small signal model will be different. From these results, following things can be clearly identified:

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Appendix C. Small signal characteristic of SRC converter

1. Beat frequency double pole. This is a special characteristic for resonant converter [5][6]. As switching frequency changes, a double pole with frequency at the difference of switching frequency and resonant frequency will move accordingly too. Finally, when switching frequency is close enough to resonant converter, this double pole will split, one merge with low frequency pole formed by output cap and load, one move to higher frequency.

Figure C.2 Bode plot of control to output transfer function of Series Resonant Converter 290

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Appendix C. Small signal characteristic of SRC converter

2. Beat frequency dynamic. Since the low frequency gain is proportional to the slope of DC characteristic of series resonant converter. When the operation frequency moves close to the resonant frequency, the slope gets flat and low frequency gain drops. When switching frequency equals to resonant frequency, the gain will be zero. As can be clearly seen on the graph, when switching frequency is close to resonant frequency, the control to output gain will be very low. A gap can be observed on the graph. 3. The phase has a 180-degree jump around resonant frequency. This is because of the change of the DC characteristic slope. Switching frequency lower than resonant frequency, with increasing switching frequency, gain will increase, so the phase delay at DC will be zero. When the switching frequency is higher than resonant frequency, as switching frequency increases, gain will decrease, which will give 180 degree at DC. 4. Low frequency pole, which is caused by the output capacitor and load. With lighter load, this pole will move to lower frequency. There should have an ESR zero if ESR of the output capacitor is considered. Here in this simulation, ESR is neglected. From above simulation results, the small signal characteristic of a series resonant converter is derived. Beat frequency double pole and beat frequency dynamic are observed. Compare with results reported in [6], a very good match is achieved. From this result, we can be more confident with the method. These

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Appendix C. Small signal characteristic of SRC converter

results also will be used as a reference to compare with LLC resonant converter since it is very similar to SRC in some operating region.

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Appendix D. LLC resonant converter model for extended describing function analysis

Appendix D. LLC converter model for EDF analysis In this part, the model file and software package for extended describing function analysis will be listed. This software is written in MATLAB. This appendix is divided into two parts. First part is the process for building the model file for extended describing function analysis. In the second part, the LLC model file for extended describing function analysis at listed. The software package for extended describing function could be found in dissertation of Dr. Eric X. Yang. The version used for the analysis is MATLAB 5.0.

D.1. Process of building LLC circuit model for EDF analysis To build the model of the converter for describing function analysis, first the operation of the converter need to be understood clearly. The operating stages in each switching cycle need to be identified. For each operating stage, the state space needs to be derived. Base on this information, the model file could be build. In this part, the operating modes of LLC converter at full load condition will be analyzed in region 1 and region 2. Region 3 is eliminated because of ZCS operation. At light load condition, the converter might run into DCM. Since DCM

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Appendix D. LLC resonant converter model for extended describing function analysis

operation will introduce many more operating modes, it is not included in this model. The circuit and notifications are shown in Figure D.1.

Figure D.1 Circuit diagram and notification for extended describing function analysis

In this circuit, there are four passive components: Lr, Cr, Lm and Co. Four states could be chosen for each components as: ILr, ILm, VCr, and VCo. But look at the topology; the current through output is the difference of two states, ILr and ILm. Also, the converter changes stage if ILr-ILm changes sign as will shown later. For simplification, the states were chosen as: ILr-ILm, ILm, VCr, and VCo. As shown in the circuit, the input variables are Vin and Io. The output variables are Iin and Vo. With these variables the state equations in each region could be derived in the form of: x& = Ax + Bu y = Cx + Du

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Appendix D. LLC resonant converter model for extended describing function analysis

x = (iLr − iLm

where u = (vin y = (vo

D.1.1.

′ io ) ′ iin )

iLm

vCr

′ vCf )

.

Model of LLC resonant converter in region 1

The simulation waveforms of LLC resonant converter in region 1 are shown in Figure D.2. The operation in this region could be divided into 4 different modes as shown in the diagram. The simplified topology in each mode and the condition for transferring from one mode to next mode is shown in Figure D.3.

Figure D.2 Simulation waveform of LLC converter in region 1

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Appendix D. LLC resonant converter model for extended describing function analysis

Figure D.3 Topology modes and progressing condition for region 1

The state equations for each mode are shown as following. Mode 1  Rs + R R − Lr − Lm  R  Lm A= 1   Cr  k −  Cf 

−

Rs Lr

−

1 Lr

0

0

1 Cr

0

0

0

k k  + Lr Lm  k  −  Lm   0  k  −  R ⋅ Cf 

296

R = Ro // Rc R k= R + Rc . Where

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1  Lr  0 B= 0 0 

Appendix D. LLC resonant converter model for extended describing function analysis

R R  + Lr Lm  R  −  Lm  0  k  Cf 

− R 0 0 k  C=   1 1 0 0 0 R  D=  0 0  Mode 2  Rs + R R − Lr − Lm  R  Lm A= 1   Cr  k  Cf  1  Lr  0 B= 0 0 

−

−

Rs Lr

−

1 Lr

0

0

1 Cr

0

0

0

−

k k  − Lr Lm   k  Lm   0   k −  R ⋅ Cf 

R R  − Lr Lm   R  Lm  0  k   Cf

R 0 0 k  C=   1 1 0 0

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Appendix D. LLC resonant converter model for extended describing function analysis

0 R  D=  0 0  Mode 3  Rs + R R − Lr − Lm  R  Lm A= 1   Cr  k  Cf   1 − Lr   0 B=  0  0 

−

Rs Lr

−

−

1 Lr

0

0

1 Cr

0

0

0

−

k k  − Lr Lm   k  Lm   0   k −  R ⋅ Cf 

R R  − Lr Lm   R  Lm  0  k   Cf

 R 0 0 k C=  − 1 − 1 0 0  0 R  D=  0 0  Mode 4

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Appendix D. LLC resonant converter model for extended describing function analysis

 Rs + R R − Lr − Lm  R  Lm A= 1   Cr  k −  Cf   1 − Lr   0 B=  0  0 

−

Rs Lr

−

1 Lr

0

0

1 Cr

0

0

0

k k  + Lr Lm  k   − Lm   0  k  −  R ⋅ Cf 

R R  + Lr Lm  R  −  Lm  0  k  Cf 

− R 0 0 k  C=   − 1 − 1 0 0 0 R  D=  0 0 

D.1.2.

Model of LLC resonant converter in region 2

The simulation waveforms of LLC resonant converter in region 2 are shown in Figure D.4. The operation in this region also could be divided into 4 different modes as shown in the diagram. The simplified topology in each mode and the condition for transferring from one mode to next mode is shown in Figure D.5.

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Appendix D. LLC resonant converter model for extended describing function analysis

Figure D.4 Simulation waveform of LLC converter in region 2

Figure D.5 Topology modes and progressing condition for region 2

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Appendix D. LLC resonant converter model for extended describing function analysis

The state equations for each mode are shown as following. Mode 1  Rs + R R − Lr − Lm  R  Lm A= 1   Cr  k  Cf  1  Lr  0 B= 0 0 

−

−

Rs Lr

−

1 Lr

0

0

1 Cr

0

0

0

−

k k  − Lr Lm   k  Lm   0   k −  R ⋅ Cf 

R R  − Lr Lm   R  Lm  0  k   Cf

R 0 0 k  C=   1 1 0 0 0 R  D=  0 0  Mode 2

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Appendix D. LLC resonant converter model for extended describing function analysis

0 0 Rs  0 − Lm + Lr  1 A = 0 Cr  0 0   0  1  B =  Lr +0 Lm   0 

  0   0   k  − R ⋅ Cf 

0 1 − Lm + Lr

0

0 0

0   0  0  k   Cf 

0 0 0 k  C=  1 1 0 0  0 R  D=  0 0  Mode 3  Rs + R R − Lr − Lm  R  Lm A= 1   Cr  k −  Cf 

−

Rs Lr

−

1 Lr

0

0

1 Cr

0

0

0

k k  + Lr Lm  k  −  Lm   0  k  −  R ⋅ Cf 

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 1 − Lr   0 B=  0  0 

Appendix D. LLC resonant converter model for extended describing function analysis

R R  + Lr Lm  R  −  Lm  0  k  Cf 

− R 0 0 k  C=   − 1 − 1 0 0 0 R  D=  0 0  Mode 4 0 0 Rs  0 − Lm + Lr  1 A = 0 Cr  0 0  0  1  − Lr + Lm B= 0  0  

0 1 − Lm + Lr 0 0

  0   0   k  − R ⋅ Cf  0

0   0  0  k   Cf 

 0 0 0 k C=  − 1 − 1 0 0 

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Appendix D. LLC resonant converter model for extended describing function analysis

0 R  D=  0 0  With above information, the model file could be written as shown next.

D.2. LLC resonant converter model Following is the model file for LLC resonant converter. It covers the operating region 1 and 2. Region 3 is now covered since it is ZCS and not preferred for this converter. This model file only deals with the normal operation mode. During very light load condition, the converter might work into different discontinuous conduction modes. To cover these operation modes, the model file needs to be extended. The method of building the model file is demonstrated in chapter 4. %%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%% % Name: topo.m -------- LLC topology for continuous condition mode % Build by: Bo Yang, Dec. 2001 % Function: define converter circuit, operating condition, % and switching boundary condition % Input: CP ------------ Circuit Parameters % x ------------- current state vector % u ------------- current input vector % contl --------- control parameters % cur_mode ------ current topological mode % t ------------- current time % % Output: num_mode ------ # of modes in one cycle % Para ---------- Circuit Parameters % x0 ------------ initial condition % U0 ------------ given input vector % CTL ----------- Control Parameters % harm_tbl ------ harmonic table (see Chap.3) % switching ----- 1 = not cross switching boundary % -1 = cross switching boundary % A, B, C, D ---- state matrices of current mode % Ab,Bb,Cb,Db---- boundary matrices of current mode % % Calling: none % %%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%% function [RT1, RT2, RT3, RT4, RT5, RT6, Bb, Cb, Db, Fo, Zo] ... = topo(CP, x, u, contl, cur_mode, t)

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Appendix D. LLC resonant converter model for extended describing function analysis

%%%%%%%%%%%%%%%%%%% State Equation Description %%%%%%%%%%%%%%%%%% % Input: U = [Vg, Io]; % Output: Y = [Vo, Ig]; % State: X = [Ilr, Ilm, Vcr, Vcf]; %%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%%% % define # of mode and dimension of output num_mode = 4; % define harmonic table % dc 1st 2nd harm_tbl = [0 1 0 0 1 0 0 1 0 1 0 0

3rd 1 1 1 0

4th 0 0 0 0

5th 1 1 1 0

0 0 0 0

1; 1; 1; 0];

% Ilr-Ilm -- 1st state % Ilm -- 2nd state % Vcr -- 3nd state % vo - 4rd state

% define circuit parameters: [Lr; Cr; Lm; Cf; rs; rc; Qs] Para = [22.5e-6; 28e-9; 60e-6; 20e-6; 0.01; 0.01; 0.5]; % define initial condition x0 = [0; -5; -300; 190];

% [Ilr-Ilm; Ilm; Vcr; vcf]

% define Input variables U0 = [200; 0];

% [Vg, Io]

% define control variables CTL = [0.99*200e3];

% Fs; Frequency control

if nargin == 0, RT1 = num_mode; RT2 = Para; RT3 = x0; RT4 = U0; RT5 = CTL; RT6 = harm_tbl; return; elseif Lr Cr Lm Cf rs rc Qs

nargin == 6, = CP(1); = CP(2); = CP(3); = CP(4); = CP(5); = CP(6); = CP(7);

% Qs=Zo/R;

Fs = contl(1); % Some parameters Zo = sqrt(Lr/Cr); Fo = 1/(2*pi*sqrt(Lr*Cr)); R = Zo / Qs;

% Zo % Fo=200 kHz;

% Fsn = Fs / Fo; k = R / (R+rc); r = k * rc; Ts = 1/Fs; % define switching boundary conditions if cur_mode == 1, Si=1; St=1;

% rectifier polarity % active bridge polarity

% set crossing boundary flag

305

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Appendix D. LLC resonant converter model for extended describing function analysis

if x(1) < 0.0 , switching = -1; else switching = 1; end % define piecewise linear state A = [(-rs-r)/Lr, -rs/Lr, r/Lm, 0, 1/Cr, 1/Cr, k/Cf, 0, B = [1/Lr, -r/Lr; 0, r/Lm; 0, 0; 0, k/Cf]; C = [r, 0, 1, 1, D = [0, r; 0, 0];

equations -1/Lr, 0, 0, 0,

-k/Lr; k/Lm; 0; -k/R/Cf];

0, 0,

k; 0];

% switching boundary condition: % Ab * x + Bb * u + Cb * t + Db < 0 Ab = [1, 0, 0, 0]; Bb = [0, 0]; Cb = 0; Db = 0; elseif cur_mode == 2, Si=-1; St=1; if t > 0.5 * Ts; switching = -1; else switching = 1; end % define piecewise linear state A = [0, 0, 0, -rs/(Lm+Lr) 0, 1/Cr, 0, 0, 0, B = [0, 0; 1/(Lm+Lr) 0; 0, 0; 0, k/Cf]; C = [0, 0, 1, 1, D = [0, r; 0, 0]; Ab Bb Cb Db

= = = =

equations 0, 0; -1/(Lm+Lr), 0; 0; 0, -k/R/Cf];

0, 0,

[0, 0, 0, 0]; [0, 0]; 0; 0;

elseif cur_mode == 3, Si=-1; St=-1; if x(1) > 0, switching = -1; else, switching = 1;

306

k; 0];

Bo Yang

Appendix D. LLC resonant converter model for extended describing function analysis

end, % define piecewise linear state A = [(-rs-r)/Lr, -rs/Lr, r/Lm, 0, 1/Cr, 1/Cr, -k/Cf, 0, B = [-1/Lr, r/Lr; 0, -r/Lm; 0, 0; 0, k/Cf]; C = [-r, 0, -1, -1, D = [0, r; 0, 0]; Ab Bb Cb Db

= = = =

equations -1/Lr, 0, 0, 0,

0, 0,

k/Lr; -k/Lm; 0; -k/R/Cf];

k; 0];

[-1, 0, 0, 0]; [0, 0]; 0; 0;

elseif cur_mode == 4, Si=1; St=-1; if t > Ts, switching = -1; else switching = 1; end % define piecewise linear state equations A = [0, 0, 0, 0; 0, -rs/(Lm+Lr) -1/(Lm+Lr), 0; 00, 1/Cr, 0, 0; 0, 0, 0, -k/R/Cf]; B = [0, 0; -1/(Lm+Lr), 0; 0, 0; 0, k/Cf]; C = [0, 0, 0, k; -1, -1, 0, 0]; D = [0, r; 0, 0]; Ab Bb Cb Db

= = = =

[0, 0, 0, 0]; [0, 0]; 0; 0;

end

RT1 RT2 RT3 RT4 RT5 RT6

= = = = = =

switching; A; B; C; D; Ab;

return; else, error('ERROR USE OF TOPO()!!'); end

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Reference

Reference A. Distributed power system and PWM topologies

[A-1] IBM Corporation, 1999 IBM Power Technology Symposium, Theme: DC/DC Conversion, September 1999, Research Triangle Park, NC. [A-2] Intel Corporation, Intel Power Supply Technology Symposium, June 1998, Dupont, WA. [A-3] Intel Corporation, Intel Power Supply Technology Symposium, September 2000. [A-4] C.C. Heath, “The Market For Distributed Power Systems,” Proc. IEEE APEC '91, 1991, 10-15 March 1991 pp. 225 –229. [A-5] W.A. Tabisz, M.M. Jovanovic, and F.C. Lee, “Present And Future Of Distributed Power Systems,” Proc. IEEE APEC '92, 1992, pp. 11 –18. [A-6] F.C. Lee, P. Barbosa, P. Xu; J. Zhang; B. Yang; Canales, F., “Topologies And Design Considerations For Distributed Power System Applications,” Proceedings of the IEEE, Volume: 89 Issue: 6, June 2001 Page(s): 939 – 950. [A-7] Mazumder, Sudip K. Shenai, Krishna, “On The Reliability Of Distributed Power Systems: A Macro- To Micro- Level Overview Using A Parallel Dc-Dc Converter,” Proc. IEEE PESC, 2002, pp. 809-814. [A-8] Hua, G.C., Tabisz, W.A., Leu, C.S., Dai, N., Watson, R., Lee, F.C., “Development Of A DC Distributed Power System,” Proc. IEEE APEC '94. 1994,pp. 763 -769 vol.2. [A-9] G.S. Leu, G. Hua, and F.C. Lee, “Comparison of Forward Circuit Topologies with various reset scheme,” VPEC 9th Seminar, 1991. [A-10] Min Chen, Dehong Xu, M. Matsui, “Study On Magnetizing Inductance Of High Frequency Transformer In The Two Transistor Forward Converter,” Proc. Power Conversion Conference, 2002, pp. 597-602. [A-11] Jianping Xu, Xiaohong Cao, Qianchao Luo, “An Improved Two Transistor Forward Converter,” Proc. IEEE PEDS, 1999, pp. 225-228.

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[A-12] W. Chen, F.C. Lee, M.M. Jovanovic, J.A. Sabate, “A Comparative Study Of A Class Of Full Bridge Zero Voltage Switched PWM Converters,” Proc. IEEE APEC, 1995, pp. 893-899. [A-13] A.W. Lotfi, Q. Chen, F.C. Lee, “A Nonlinear Optimization Tool For The Full Bridge Zero Voltage Switched DC-DC Converter,” Proc. IEEE PESC ‘92, 1992, pp. 1301-1309. [A-14] Xinbo Ruan; Yangguang Yan, “Soft-Switching Techniques For PWM Full Bridge Converters,” Proc. IEEE PESC ’00, 2000, pp. 634 -639 vol.2. [A-15] D.B. Dalal, F.S. Tsai, “A 48V, 1.5kw, Front-End Zero Voltage Switches PWM Converter With Lossless Active Snubbers For Output Rectifiers,” Proc. IEEE APEC 93, 1993, pp. 722-728. [A-16] J.G. Cho, J.A. Sabate, G. Hua, and F.C. Lee, “Zero Voltage And Zero Current Switching Full Bridge PWM Converter For High Power Applications,” Proc. IEEE PESC ’94, 1994. [A-17] J.H. Liang, Po-chueh Wang, Kuo-chien Huang, Cern-Lin Chen, Yi-Hsin Leu, Tsuo-Min chen, “ Design Optimization For Asymmetrical Half Bridge Converters,” Proc. IEEE APEC ’01, 2001, pp. 697-702, vol.2. [A-18] Y. Leu, C. Chen, “Analysis and Design of Two-Transformer Asymmetrical Half-Bridge Converter,” Proc. IEEE PESC ’02, 2002, pp. 943-948. [A-19] S. Korotkov, V. Meleshin, R. Miftahutdinov, S. Fraidlin, “Soft Switched Asymmetrical Half Bridge DC/DC Converter: Steady State Analysis. An Analysis of Switching Process,” Proc. Telescon ’97,. 1997, pp. 177-184. [A-20] L. Krupskiy, V. Meleshine, A. Nemchinov, “Unified Model of the Asymmetrical Half Bridge for Three Important Topological Variations,” Proc. IEEE INTELEC ’99, 1999, pp.8. [A-21] L. Krupskiy, V. Meleshine, A. Nemchinov, “Small Signal Modeling of Soft Switched Asymmetrical Half Bridge DC/DC Converter,” Proc. IEEE APEC ’95, 1995, pp. 707-711. [A-22] W. Chen, P. Xu, F.C. Lee, “The Optimization Of Asymmetrical Half Bridge Converter,” Proc. IEEE APEC ’01, 2001, pp. 603-707, vol.2. [A-23] P. Wong, B. Yang, P. Xu and F.C. Lee, “Quasi Square Wave Rectifier For Front End DC/DC Converters,” Proc. IEEE PESC ’00, 2000, pp. 10531057, vol.2.

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[A-24] B. Yang, P. Xu, and F.C. Lee, “Range Winding For Wide Input Range Front End DC/DC Converter,” Proc. IEEE APEC, 2001, pp. 476-479. [A-25] G. Hua, and F.C. Lee, “An Overview of Soft-Switching Techniques for PWM Converters,” Proc. IPEMC ’94, 1994.

B. Resonant converter

[B-1] R. Oruganti and F.C. Lee, “Resonant Power Processors, Part 2: Methods of Control,” IEEE Trans. on Industrial Application, 1985. [B-2] R. Severns, “Topologies For Three Element Resonant Converters,” Proc. IEEE APEC ’90, 1990, pp. 712-722. [B-3] Vatche Vorperian, Analysis Of Resonant Converters, Dissertation, California Institute of Technology, 1984. [B-4] R. Farrington, M.M. Jovanovic, and F.C. Lee, “Analysis of Reactive Power in Resonant Converters,” Proc. IEEE PESC ’92, 1992. [B-5] P. Calderira, R. Liu, D. Dalal, W.J. Gu, “Comparison of EMI Performance of PWM and Resonant Power Converters,” Proc. IEEE PESC ’93, 1993, pp. 134-140. [B-6] T. Higashi, H. Tsuruta, M. Nakahara, “Comparison of Noise Characteristics for Resonant and PWM Flyback Converters,” Proc. IEEE PESC ’98, 1998, pp. 689-695. [B-7] K.H. Liu, “Resonant Switches Topologies and Characteristics,” Proc. IEEE PESC’85, 1985, pp. 106-116. [B-8] R. Oruganti and F.C. Lee, “Effect of Parasitic Losses on the Performance of Series Resonant Converters,” Proc. IEEE IAS, ’85, 1985. [B-9] R. Oruganti, J. Yang, and F.C. Lee, “Implementation of Optimal Trajectory Control of Series Resonant Converters,” Proc. IEEE PESC ’87, 1987. [B-10] A.K.S. Bhat, “Analysis and Design of a Modified Series Resonant Converter,” IEEE Trans. on Power Electronics, 1993, pp. 423-430. [B-11] J.T. Yang and F.C. Lee, “Computer Aided Design and Analysis of Series Resonant Converters,” Proc. IEEE IAS ’87, 1987. 310

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[B-12] V. Vorperian and S. Cuk, “A Complete DC Analysis of the Series Resonant Converter,” Proc. IEEE PESC’82, 1982. [B-13] F.S. Tsai, and F.C. Lee, “A Complete DC Characterization of a ConstantFrequency, Clamped-Mode, Series Resonant Converter,” Proc IEEE PESC’88, 1988. [B-14] R. Oruganti, J. Yang, and F.C. Lee, “State Plane Analysis of Parallel Resonant Converters,” Proc. IEEE PESC ’85, 1985. [B-15] R. Liu, I. Batarseh, C.Q. Lee, “Comparison of Capacitively and Inductively Coupled Parallel Resonant Converters,” IEEE Trans. on Power Electronics, 1993, pp. 445-454, vol.8, issue 4. [B-16] M. Emsermann, “An Approximate Steady State and Small Signal Analysis of the Parallel Resonant Converter Running Above Resonance,” Proc. Power Electronics and Variable Speed Drives ’91, 1991, pp. 9-14. [B-17] Y.G. Kang, A.K. Upadhyay, D. Stephens, “Analysis and Design of a Half Bridge Parallel Resonant Converter Operating Above Resonance,” Proc. IEEE IAS ’98, 1998, pp. 827-836. [B-18] Robert L. Steigerwald, “A Comparison of Half Bridge Resonant Converter Topologies,” IEEE Trans. on Power Electronics, 1988, pp. 174-182. [B-19] M. Zaki, A. Bonsall, I. Batarseh, “Performance Characteristics for the Series Parallel Resonant Converter,” Proc. Southcon ’94, 1994, pp.573577. [B-20] A.K.S. Bhat, “Analysis, Optimization and Design of a Series Parallel Resonant Converter,” Proc. IEEE APEC ’90, 1990, pp. 155-164. [B-21] H. Jiang, G. Maggetto, P. Lataire, “Steady State Analysis of the Series Resonant DC/DC Converter in Conjunction with Loosely Coupled Transformer Above Resonance Operation,” IEEE Trans. on Power Electronics, 1999, pp. 469-480. [B-22] R. Liu, C.Q. Lee, and A.K. Upadhyay, “A Multi Output LLC-Type Parallel Resonant Converter,” IEEE Trans. on Aerospace and Electronic Systems, 1992, pp. 697-707, vol.28, issue 3. [B-23] A.K.S. Bhat, “Analysis and Design of a Series Parallel Resonant Converter With Capacitive Output Filter,” Proc. IEEE IAS ’90, 1990, pp. 1308-1314.

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[B-24] R. Liu, and C.Q. Lee, “Series Resonant Converter with Third-Order Commutation Network,” IEEE Trans. on Power Electronics, 1992, pp. 462-467, vol.7, issue 3. [B-25] J.F. Lazar, R. Martinelli, “Steady State Analysis of the LLC Series Resonant Converter,” Proc. IEEE APEC’01, 2001, pp.728-735. [B-26] R. Elferich, T. Duerbaum, “A New Load Resonant Dual Output Converter,” Proc. IEEE PESC’02, 2002, pp.1319-1324. [B-27] B. Yang, F.C. Lee, A.J. Zhang and G. Huang, “LLC Resonant Converter For Front End DC/DC Conversion,” Proc. IEEE APEC, 2002, pp. 11081112 vol.2.

C. Integrated Magnetic

[C-1] P. -L. Wong, P. Xu, B. Yang and F. C. Lee, “Performance Improvements of Interleaving VRMs with Coupling Inductors,” Proc. CPES Annual Seminar, 2000, pp. 317-324. [C-2] B. Mohandes, “Integrated PC Board Transformers Improve High Frequency PWM Converter Performance,” PCIM Magazine, July 1994. [C-3] I. Jitaru and A. Ivascu, “Increasing the Utilization of the Transformer’s Magnetic Core by Using Quasi-Integrated Magnetics,” Proc. HFPC Conf., 1996, pp. 238-252. [C-4] C. Peng, M. Hannigan, O. Seiersen, “A New Efficient High Frequency Rectifier Circuit”, Proc. HFPC Conf., 1991, pp. 236-243. [C-5] P. Xu, Q. Wu, P.-L. Wong and F. C. Lee, “A Novel Integrated Current Doubler Rectifier,” Proc. IEEE APEC, 2000, pp. 735-740. [C-6] S. Cuk, “New Magnetic Structures for Switching Converters,” IEEE Transaction On Magnetics, March 1983, pp. 75-83. [C-7] S. Cuk, L. Stevanovic and E. Santi, “Integrated Magnetic Design with Flat Low Profile Core,” Proc. HFPC Conf., 1990. [C-8] I. W. Hofsajer, J. D. van Wyk and J. A. Ferreira, “Volume Considerations of Planar Integrated Components,” Proc. IEEE PESC, 1999, pp. 741-745.

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[C-9] D. Y. Chen, “Comparison of High Frequency Magnetic Core Loss under Two Different Driving Conditions: A Sinusoidal Voltage and a SquareWave,” Proc. IEEE PESC, 1978, pp. 237-241. [C-10] N. Dai and F. C. Lee, “Edge Effect Analysis in a High-Frequency Transformer,” Proc. IEEE PESC, 1994, pp. 850-855. [C-11] W. M. Chew and P. D. Evans, “High Frequency Inductor Design Concept,” Proc. IEEE PESC, 1991, pp. 673-678. [C-12] K. D. T. Ngo and M. H. Kuo, “Effects of Air Gaps on Winding Loss in High-Frequency Planar Magnetics,” Proc. IEEE PESC, 1988, pp. 11121119. [C-13] L. Ye, R. Wolf and F. C. Lee, “Some Issues of Loss Measurement and Modeling for Planar Inductors,” Philips Fellowship Report, VPEC, September 1997. [C-14] N. Dai, Modeling, Analysis, and Design of High-Frequency, High-Density, Low-Profile Transformers, Dissertation, Virginia Tech, Blacksburg, VA, 1996. [C-15] M. T. Zhang, M. M. Jovanovic and F. C. Lee, “Analysis, Design and Evaluation of Forward Converter with Distributed Magnetics – Interleaving and Transformer Paralleling,” Proc. IEEE APEC, 1995, pp. 315-321. [C-16] R. Prieto, J. A. Cobos, O. Garcia and J. Uceda. “Interleaving Techniques in Magnetic Components,” Proc. IEEE APEC, 1997, pp. 931-936. [C-17] Rengang Chen; Canales, F.; Bo Yang; van Wyk, J.D., “Volumetric Optimal Design Of Passive Integrated Power Electronic Module (IPEM) For Distributed Power System (DPS) Front-End DC/DC Converter,” Proc. IEEE IAS, 13-18 Oct. 2002 pp. 1758 -1765 vol.3. [C-18] Rengang Chen, J.T. Strydom, J.D. Van Wyk, “Design Of Planar Integrated Passive Module For Zero Voltage Switched Asymmetrical Half Bridge PWM Converter,” Proc. IEEE IAS, 2001, pp. 2232-2237 vol.4. [C-19] B. Yang, Rengang Chen, and F.C. Lee, “Integrated Magnetic For LLC Resonant Converter,” Proc. IEEE APEC, 2002, pp. 346-351, vol.1. [C-20] Chen, Rengang. Yang, Bo. Canales, F. Barbosa, P. Van Wyk, J D. Lee, F C. “Integration Of Electromagnetic Passive Components In DPS Front-

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Bo Yang

Reference

End DC/DC Converter - A Comparative Study Of Different Integration Steps,” Proc. IEEE APEC. 2003. p 1137-1142. [C-21] P. -L. Wong, Q. Wu, P. Xu, B. Yang and F. C. Lee, “Investigating Coupling Inductor in Interleaving QSW VRM,” Proc. IEEE APEC Conf., 2000, pp. 973-978. [C-22] W. –J Gu, R. Liu, “A Study of Volume and Weight vs. Frequency for High-Frequency Transformers,” Proc. IEEE PESC ’93, 1993, pp. 11231129. [C-23] H.A. Kojori, J.D. Lavers, S.B. Dewan, “State Plane Analysis of a Resonant DC-DC Converter Incorporating Integrated Magnetic,” IEEE Trans. on Magnetics, 1988, pp. 2898-2900. [C-24] A. Kats, G. Ivensky, S. Ben-Yaakov, “Application of Integrated Mangetics in Resonant Converters,” Proc. IEEE APEC’97, 1997, pp. 925930.

D. Small Signal Modeling [B-1]

G. W. Wester and R. D. Middlebrook, “Low-Frequency Characterization of Switched DC-to-DC Converters,” Proc. IEEE PESC, 1972, pp. 9-20.

[B-2]

R. D. Middlebrook, “Measurement of Loop Gain in Feedback Systems,” Int. J. Electronics, 1975, pp. 485-512.

[B-3]

R. D. Middlebrook and S. Cuk, “A General Unified Approach to Modeling Switching-Converter Power Stages,” Proc. IEEE PESC, 1976.

[B-4]

R. Tymerski and V. Vorperian, “Generation, Classification and Analysis of Switched-Mode DC-to-DC Converters by the Use of Converter Cells,” Proc. INTELEC, 1986, pp. 181-195.

[B-5]

V. Vorperian, “Simplified Analysis of PWM Converters Using Model of PWM Switch, Part I: Continuous Conduction Mode,” IEEE Trans. on Aerospace and Electronic Systems, 1990, pp. 490-496.

[B-6]

V. Vorperian, “High Q Approximate Small Signal Analysis of Resonant Converters,” VPEC Seminar 1984.

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Reference

[B-7]

V. Vorperian, “Simplified Analysis of PWM Converters Using Model of PWM Switch, Part II: Discontinuous Conduction Mode,” IEEE Trans. on Aerospace and Electronic Systems, 1990, pp. 497-505.

[B-8]

J. O. Groves and F. C. Lee, “Small Signal Analysis of Systems with Periodic Operating Trajectories,” Proc. VPEC Annual Seminar, 1988, pp. 224-235.

[B-9]

J. O. Groves, “Small-Signal Analysis Using Harmonic Balance Methods,” Proc. IEEE PESC, 1991, pp. 74–79.

[B-10]

E. X. Yang, “Extended Describing Function Method for Small-Signal Modeling of Switching Power Circuit,” Proc. VPEC Annual Seminar, 1994, pp.87-96.

[B-11]

E. X. Yang, F. C. Lee and M. Jovanovic, “Small-Signal Modeling of Series and Parallel Resonant Converters,” Proc. IEEE APEC, 1992, pp. 785-792.

[B-12]

Eric X. Yang, Extended Describing Function Method for Small-Signal Modeling of Resonant and Multi-Resonant Converters, Dissertation, Virginia Tech, Blacksburg, VA, February 1994.

[B-13]

R.C. Wong and J. O. Groves, “An Automated Small-Signal FrequencyDomain Analyzer for General Periodic-Operating Systems as Obtained via Time-Domain Simulation,” Proc. IEEE PESC, 1995, pp. 801-808.

E. General Textbooks

[E-1] N. Mohan, T. Undeland and W. Robbins, Power Electronics, 1995. [E-2] Abraham I. Pressman, Switching Power Supply Design, 1991. [E-3] Rudolf P. Severns, Modern DC-To-DC Switchmode Power Converter Circuits, 1985. [E-4] J.K. Watson, Applications Of Magnetism, 1985

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Bo Yang

Vita

Vita The author, Bo Yang, was born in Linwu, Ningxia, P.R. China in 1972. He received B.S and M.S degrees in Electrical Engineering from Tsinghua University, Beijing, China respectively. In fall 1997, the author joined the Virginia Power Electronics Center (VPEC) - now the Center for Power Electronics Systems (CPES) - at Virginia Polytechnic Institute and State University, Blacksburg, Virginia. His research interests include power converter simulation and modeling, magnetic design and modeling, DC/DC converters, and voltage regulator modules (VRMs). The author is a member of the IEEE Power Electronics Society.

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