All Time Favorite Electronic Projects
Commodore Computer Book...
"" Delton T.Horn's
Ail-Time Favorite Electronic Projects
DeltonT. Horn's All-Time Favorite Electronic Projects
All-Time Favorite Electronic Projects DeltonT.Horn
TAB BOOKS Blue Ridge Summit, PA
FIRST EDITION FOURTH PRINTING ©1989 by TAB Books. TAB Books is a division of McGraw-Hill, Inc.
Printed in the United States of America. All rights reserved. The publisher takes no responsibility for the use of any of the materials or methods described in this book, nor for the products thereof.
Library of Congress Cataloglng-in-Publication Data Horn, Delton T.
[All-time favorite electronic projects] Delton T. Horn's all-time favorite electronic projects / by Delton T. Horn p.
Includes index. ISBN 0-8306-3105-4 (pbk.)
1. Electronics—Amateurs1 manuals. -time favorite electronic projects. TK9965.H639 1988
II. Title: All
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Contents Editor's Preface PART I: For the Home
Car Thief Alarm
Electric Motor Speed Controller
6 Audio Amplifiers
Tape Player Amplifier
Hi-Fi Tone Controls
PART U: For the Shop
10 Constant Cunent or Constant Voltage Source
11 Digital-to-Analog Converter
12 Logic Probe
13 Digital Capacitance Meter
14 Digital Frequency Meter
15 Multiple-Output Power Supply
16 Dc Voltmeter
and has worked in a variety of fields as an electronics technician. He is now occupied as a full-time technical author. All 16 of these projects were extracted from six of his best projects books currently published by TAB. These books are: 117Practical IC Projects You Can Build (TAB book #2645), Using Integrated Circuit Logic Devices (TAB book #1645), Amplifiers Simplified (TAB book #2885), Transistor Circuit Design with Experiments (TAB book #1875), How to Design Op Amp Circuits with Projects &
Experiments (TAB book #1765), and Designing IC Circuits (TAB book #1925). Numerous other tides covering all aspects of electronics are available from TAB. Also available is a new line of electronics engineering and design titles by Mr. Horn and by many other authors. The variety of projects in this book will appeal to both the beginner and the more experienced electronic hobbyist. However, the emphasis is on practicality. This text is divided into two parts. Part I is For the Home. Included are gadgets like setting up an intercom or building your own digital clock. Also contained in this
section are projects you can construct to create or build upon audio equipment, such as amplifiers and tone controls. Part II discusses items For the Shop. Described are a variety
ply. When completed and added to your current equipment, these vii
handy devices will make your shop an easier and more enjoyable place to work. It is sincerely hoped you will enjoy these projects while simultaneously learning and producing a useftil, working item for your
home or shop.
For the Home
PROJECT 1: Intercom The TCA830S is a powerful, inexpensive op amp IC that makes it a particularly attractive choice for intercoms because the circuit can be built with a minimum number of components. Many other op amps do not produce the power required for loudspeaker operation with out the addition of a further stage of transistor amplification. The basic circuit is contained at the main station while the distant station merely comprises a loudspeaker and a calling switch. The two stations are connected by a 3-wire flex. The circuit is shown in Fig. 1-1. The TCA830S requires a heatsink and is fitted with tabs. A printed circuit is recommended, incorporating two 1-inch (25 mm) squares of copper to which the IC tabs can be soldered for the heatsink. Component positioning is not critical because the circuit handles only audio frequencies. The transformer (T) has a 50:1 turns ratio and is used as a stepdown transformer between the IC and speaker(s); it also works as a step-up transformer between the speakers and IC in the reverse mode. In other words, the transformer coil with the larger number of turns is connected to pin 8 on the IC. Instead of purchasing this transformer ready-made, it can be wound on a stack of standard transformer core laminates 0.35 mm thick, giving a core crosssection of 22.5 mm2. Windings are 600 turns of 0.2 mm (36 s.w.g.) and 300 turns of 0.06 mm (46 s.w.g.) enameled copper wire. The purpose of the transformer is to enable standard 40 to 16Q loudspeakers to be used both as microphones and speakers. These
Table 1-1. Parts List for Project 1: Intercom. IC Rl
R2 Cl C2 C3 T SPKRS SI S2
TCA8305 op amp 20KQ(remote station) 29 Q lOOjtF, electrolytic, 3V 0.1pF 10
1 / stations
i REMOTE STATION
Fig. 1-1. Intercom circuit using the TCA830S integrated circuit. This IC is powerful enough to operate fairly large loudspeakers. Component values are given in the text.
speakers can be of any size, bearing in mind that the maximum power output of the circuit is of the order of 2 watts on a 12-volt supply. The intercom circuit will work on any battery voltage down to 6 volts, 9 or 12 volts being recommended for general operation.
Car Thief Alarm This circuit is by Siemens and is based around their TDB0556A dual timer IC. See Fig. 2-1. The first timing circuit of this device is used as a bistable multivibrator with the circuit activated by switch SI. Output level remains at zero, set by the voltage applied to the thresh old input pin 2 until one of the alarm contact switches is closed,
causing Cl to discharge.
"Press-for off' alarm switches can be fitted to the doors, bonnet and boot lid, so arranged that opening a door or lid completes that switch contact. This produces an output signal for about 8 seconds, pulling in the relay after an initial delay of about 4 seconds. The horn circuit is completed by the relay contacts, so the horn will sound for 8 seconds. After this, the relay drops out (shutting off the horn) until capacitor Cl charges up again. This takes about 3 seconds, when the relay pulls in once more and the horn sounds again. This varying signal of 8 seconds horn on, 3 seconds horn off, is repeated until switch SI is turned off (or the battery is dead). This type of alarm signal commands more attention than a continuous sounding alarm.
Fig. 2-2. Circuit design for a car theft alarm.
TOB 3 0556A 11
Switching Relay, type K V 23033
82k h R1 L
Push button for the horn
8.5 to 15V
Electric Motor Speed Controller
such as those used in portable cassette players, movie cameras, models, and toys. The object is to govern the motor so that it runs at a constant speed, independent of variations in battery supply volt age and load on the motor. The TDA1151 is selected for the following circuits, having a maximum rating of 20 volts (which covers most model and other small dc motors), with an output current of up to 800 milliamps. It is a flat rectangular plastic package with three leads emerging from one end, and it comprises 18 transistors, 4 diodes and 7 resistors in a linear integrated circuit. In its simplest application, it is used with a potentiometer (Rg) acting as a speed regulation resistance (and by which the actual motor speed is adjusted); and a torque control resistor (R,) which provides automatic regulation against load on the motor. Both of these resistors are bridged by capacitors, although C2 can be omitted (see
Fig. 3-1). Component values shown in Table 3-1 are suitable for a
6- to 12-volt supply.
A slightly different circuit is shown in Fig. 3-2, using a TCA600/900 or TCA610/910 integrated circuit. These have maximum voltage ratings of 14 and 20 volts respectively and maximum current ratings of 400 milliamps for starting, but only 140 milliamps for continuous running.
Devices of this type work on the principle of providing a con stant output voltage to the motor independent of variations in sup-
I C2 I
Fig. 3-2. 77« TDA1151 linear integrated circuit used as a speed regulator far a small dc electric motor.
ply voltage, the value of this voltage being set by adjustment of R,. At the same time, the device can generate a negative output resis tance to compensate speed fluctuations due to variations in torque. This negative output resistance is equal to RT/K, where K is a con stant, depending on the parameters of the device, as demonstrated in Table 3-2. The table also shows the reference voltage (V^ and quiescent
current drain (!„) of the three ICs mentioned. Table 3-1. 1K0
Cl C2(if used)
280 0 10 pF to2/tF
Eg. 3-2. ApplkatimdmHfbrtheTCAM/610orTCAm/910nwt
50 to and an output resistance of 170 to.
Ffg. 64. Circuit for using first and second amplifiers contained in CA3035 in cascade. This circuit gives a voltage gain of about 7000 with an input of
Table 6-1. Parts list for Fig. 6-5. Rl R2
220 KQ 1.2 KQ
1KQ 4.7 KQ
Cl C2 C3 C4 C5 C6 C7
10 /tF 0.04/tF 0.22 pF 0.05 ?F
0.05 mF 50 pF
as a blocking capacitor in this case). Because amplifier 2 in this chip is directly coupled to amplifier 1 and amplifier 2 is directly coupled to amplifier 3, the use of an impedance-matching resistor applied to amplifier 2 (or amplifier 3) requires the use of a blocking capaci tor in series with the resistor.
The gain of the amplifier stage can be modified by the use of a series resistor in the input (Rl). This acts as a potential divider in conjunction with the effective input resistance of the stage so that only a proportion of the input signal is applied to the stage. In this case:
Rl Actual Voltage Gain = — Rj + Rl/A,
Input Resistance -
where R, is the input resistance of the IC. Thus, by suitable choice of Rl and Rj, both voltage gain and input resistance of an amplifier circuit can be modified to match specific requirements. It follows that if a number of different resistors are used for Rj, the circuit can be given different response (sensitivity) for a given input applied to each value of Rj by switch ing. This mode of working is useful for preamplifiers. Virtually the same circuit is used for an audio mixer, separate input channels being connected by separate series resistors (R) and commonly connected to the input. In this case, each channel has the same input resistance with an overall gain of unity.
Power Amplifiers Because amplifiers are used in one form or another in almost every electronic system, it is not surprising that a great many amplifier ICs have been developed for various applications. Some have incredibly impressive specifications. It is often surprising to see how much power can be packed into the tiny IC chip. For the most efficient and reliable performance, IC amplifiers should always be used with adequate heatsinking. In many cases, the better the heatsinking, the greater the maximum output power can be. When in doubt about how much heatsinking to use, try to err on the side of too much rather than too little. The chief disadvantage of using a heatsink that is too large is that the device would take up a little more space. The disadvantages of using an insufficient heatsink include inferior performance and possibly thermal damage to expensive components. It would be futile to attempt to compile a complete list of available amplifier ICs. Hundreds of devices are already on the market, with more appearing every month. A so-called' 'complete'' listing would be out of date before it came off the presses. This chapter looks at a few representative examples of amplifier ICs of various types. The only kind of amplifier IC that will not be discussed in this chapter is the op amp. Emphasis in this chapter is on audio amplifier ICs, because these are the most widely available and of the most interest to experimenters. 32
THE LM380 AUDIO AMPLIFIER IC
The LM380 is an audio amplifier IC that has been around for quite a few years now and still enjoys considerable popularity. Judging from the industrial and hobbyist literature, this chip appears to be the most popular amplifier device around. The LM380 is widely available, and it comes in two packaging styles, an 8-pin DIP and a 14-pin DIP. The pinout for the 8-pin version is shown in Fig. 7-1, while the 14-pin version is illustrated in Fig. 7-2. Notice that the 14-pin version does not have any additional pin functions. On both packages, only six of the pins are actually active. The remaining pins are shorted to ground and provide some
internal heatsinking. Since the 14-pin LM380 has more heatsink pins than the 8-pin version, it can handle greater amounts of power with out overheating. There are no other differences between the two package types. Without any external heatsinking, the basic LM380 can dissipate up to about 1.25 watts at room temperature. This certainly isn't bad for an amplifier less than the size of your thumbnail, but the LM380 can put out even more power with external heatsinking. For instance, if a 14-pin LM380 is mounted on a PC board with 2-ounce
foil, and the six heatsink pins are soldered to a 6-square-inch copper foil pad, the IC can produce up to about 3.7 watts at room temperature. This is an impressive, almost threefold increase in power at very little additional cost.
This chip also features an internal automatic thermal shutdown circuit that turns the amplifier off if excessive current flow causes
Inverting [T input
T]Output 5| Ground
Fig. 7-1. The LM380 amplifier IC, available in an 8-pin DIP housing.
Bypass jT 13]N.C.
E Ground (heatsinking)
Inverting input f^
T)n.c. T] Output
Fig. 7-2. Even the 14-pin DIP version has only six active pins; the extra pins are used for heatsinking.
the IC to start overheating. This feature significantly reduces the worry of short-circuit problems. As illustrated in Fig. 7-3, the LM380's internal circuitry is made up of a dozen transistors and associated components. Gain is internally fixed at 50 (34 dB). The output automatically centers itself at one half the supply voltage, effectively eliminating problems of offset drift. If a symmetrical (equal positive and negative voltages) dual-polarity power supply is used with the LM380, the output is centered around ground potential (0 volts), with no dc component to worry about. In this case, no output capacitor is needed to protect the speaker. The LM380 can be wired for the higher fidelity directcoupled output type of connection. The input stage of the LM380 is rather unusual. The input signal can either be referenced to ground or ac coupled, depending on the
particular requirements of the specific application. As you can see, this device offers considerable flexibility to the circuit designer. The inputs are internally biased with a 150 K on-chip resistance to ground. Transducers or earlier stages that are referenced to 34
Fig. 7-3. A simplified diagram of the internal circuitry in the LM380 amplifier 1C.
ground (no dc component) can be directly coupled to either the inverting or the non-inverting input. (These inputs are simflar to those found on op amps. The inverting input phase shifts the signal 180 degrees, while the non-inverting input does not phase shift the
In most applications, only one of the LM380's inputs is used. There are several possibilities for handling the unused input terminal: • Leave it floating • Short it directly to ground
• Reference it to ground through an external resistor or capacitor In many applications in which the non-inverting input is used, the inverting input is left floating (unconnected). This is fine, but it makes board layout critical. The designer must be on guard for any stray capacitances. While it is always true that stray capacitances
can lead to positive feedback, instability, and possible oscillations,
this configuration is particularly susceptible to such problems.
The LM380 audio amplifier IC is designed for use with a minimum of external components. The most basic form of a LM380-based amplifier circuit is shown in Fig. 7-4. Clearly, it would be difficult for a circuit to be much simpler than this. The only required external component is the output decoupling capacitor. As mentioned earlier, if a dual-polarity power supply is used, even this capacitor can be eliminated. The LM380 can certainly be considered complete in itself. In many practical applications, it might be desirable or even necessary to add several external components. For example, if the chip is located more than 2 or 3 inches from the power supply's fil ter capacitor, a decoupling capacitor should be mounted between the V + terminal of the LM380 and ground. Typically, the value of this capacitor is in the neighborhood of 0.1 i*F. For best results, this decoupling capacitor should be mounted as dose to the LM380's body as possible. The LM380 tends to become unstable and break into oscillations if it is used in a high-frequency (several megahertz or more) rather than audio application. Even though this IC is designated as an audio amplifier, it can function as an rf amplifier too. Adding an extra resistor and capacitor, as shown in Fig. 7-5, helps suppress parasitic oscillations in high-frequency applications. Generally, the resistor's value is very small. A typical value is 2.7 ohms. The capacitor is usually about 0.1 jiF.
Fig. 7A. The most basic circuit built around the LM380 amplifier IC
Fig. 7-5. Adding a resistor and a capacitor to the circuit of Fig. 7-4 helps minimize oscillation problems.
Because these parasitic oscillations only occur at 5 to 10 MHz, they obviously won't be of much importance in most audio applications. Even so, if the LM380 is being used in an rf-sensitive environment, such oscillations could pose a problem unless they are properly suppressed. (Note that in Fig. 7-5 and all the future diagrams that the heatsinking pins are not shown. This is for clarity in the circuit diagrams. The grounding of these pins is assumed in all cases.) Figure 7-6 shows a practical audio amplifier circuit using the LM380. A typical parts list for this project is given in Table 7-1. The input to this circuit can be provided by an inexpensive low
recorders. If a low impedance source is used, an impedance matching transformer is necessary. If this circuit is to be used with a highimpedance source, this transformer can be eliminated. The 1MQ potentiometer serves as a volume control. The fixed gain of this circuit can be increased by adding a little positive feedback.
An 8 Q speaker can be driven directly by the LM380 audio amplifier IC. The output decoupling capacitor is needed if a singlepolarity power supply is used, as shown in the diagram. The LM380 is often used in inexpensive tape recorders and
The input in this case is a ceramic cartridge. The parts list is given 37
Fig. 7-6. A practical audio amplifier circuit using the LM380.
Table 7-1. Parts list for Fig. 7-6. Component
LM380 audio amplifier IC Transformer—500 ohm:200 K
IC1 T1 C1 R1
500 pF 25 V electrolytic capacitor 1 Meg potentiometer
Pig. 7-7. The LM380 is ideal for use in inexpensive ceramic cartridge phonographs.
Table 7-2. Parts list for Fig. 7-7. Component Number
LM380 audio amplifier IC
0.047 pF capacitor
500 nF electrolytic capacitor
R1 R2 R3
25 K potentiometer 75 K resistor 10 K potentiometer
in Table 7-2. Potentiometer Rl is a simple voltage-divider volume control. Potentiometer R3 is a basic tone control, making the high frequency roll-off characteristics of the circuit manually adjustable. Most serious phonograph applications require frequency shaping to provide the standard RIAA equalization characteristic. All commercially available records are equalized according to the RIAA standards. Obviously, if the complementary frequency-response shaping isn't included in the playback, the reproduced sound won't be as good as it should be. Figure 7-8 shows a LM380-based phonograph amplifier circuit with full RIAA equalization. The parts list is given in Table 7-3.
Fig. 7-8. This phonograph amplifier includes RIAA equalization.
Table 7-3. Parts List for Fig. 7-8. Component Number
LM380 audio amplifier IC
C1 C2 C3
220 pF capacitor 0.0022 i»F capacitor
1.5 Meg resistor 2 Meg potentiometer
500 pF electrolytic capacitor
The mid-band gain can be defined with this formula: G = (Rl + 150000) /150000
The constant 150000 represents the internal resistance presented within the LM380 itself. Rl is the value of the external resistor. The corner frequency is determined by resistor Rl and capaci tor Cl, according to this formula:
In designing such a circuit, it is usually best to select Rl for the desired gain first, then rearrange the corner frequency equation to solve for Cl.
A pair of LM380 amplifiers can be put into a bridge configuration, as illustrated in Fig. 7-9. This is done to achieve more output power than could be obtained from a single amplifier. This circuit provides twice the voltage gain across the load for a given supply voltage, which increases the power handling capability by a factor of four over a single LM380.
When using the bridge configuration, caution is called for. The
output power below the theoretical quadruple level. A typical parts list for this bridged-amplifier circuit is given in Table 7-4. This has barely scratched the surface of LM380-based amplifier circuits. You can see now why this chip is such a best-seller. 40
rh Kg. 7-9. A pair oflM380s in a bridge configuration can produce greater output power.
Table 7-4. Parts List for Fig. 7-9. Component
LM380 audio amplifier IC
C1 C2, C3, C4
50 pF capacitor 0.1 pF capacitor
2 Meg potentiometer
Tape Player Amplifier The circuit shown in Fig. 8-1 is designed as a playback amplifier for
a tape deck. The standard NAB equalization curve is matched by the feedback network. The parts list is shown in Table 84.
The tape head should be selected to generate about 800 ^vat
1 kHz.The output is approximately 5 volts nns. An external volume
control potentiometer can be added to the output line. Table 8-1.
IC Rl R2 R3 R4 Cl C2
LM381 240 KQ
62 KQ 1*F
C3 1500 pF
,8-1. Playback amplifier for a tape deck.
Hi-Fi Tone Controls Tone controls fitted to domestic radios and equivalent circuits are seldom of high quality. This does not usually matter for AM reception (which can never be hi-fi); but can degrade the performance on FM reception. Similar remarks apply to the tone controls fitted to lower priced record players and tape recorders. High quality tone controls generally demand quite complex circuits. ICs enable the number of discrete components required to be substantially reduced and at the same time offer other advantages such as a high input impedance that matches a typical high impedance source. Tone control can also be combined with audio amplification in IC circuits. Figure 9-1 shows a complete circuit based around a TCA8305 integrated circuit incorporating a feedback network which attenuates the low frequencies and boosts the high frequencies. (Parts list shown in Table 9-1.) At the same time, high frequencies can be attenuated by the treble control potentiometer at the input. The volume control, also on the input side, provides "loudness control" at both high and low frequencies to compensate for the loss of sensitivity of the human ear to such frequencies (i.e., both high and low frequencies tend to sound "less loud" to the ear). A simpler circuit, using the same IC, is shown in Fig. 9-2, with parts list in Table 9-2. This has a single tone control potentiometer. The circuit provides flat response at middle frequencies (i.e., around 1 kHz), with marked boost and cut of up to ± 10 decibels at 110 44
Fig. 94, Hi-fi tone control circuit suitable for receivers, record players and tape recorders and characterized by a high input impedance. Potentiometer Rl is the treble control. Potentiometer R9 is the bass control. Potentiometer R4 is the volume control.
Table 9-1. Parts list for Fig. 9-1. Rl R2 R3
R4 R5 R6 R7 R8 R9
47 KQ log pot 10 KQ 1.8 KQ 100 KQ log pot 100 Q 15 Q 470Q 470 O 25 KQ log pot 1Q
C3 C4 C5 C6 C7 C8 C9 C10 Cll C12
820 pF 100 pF 0.1 pF 100 a»F 250 pF 100 jiF
100 pF 0.33/iF 0.22 pF 0.1 nF 1000 jtF
Fig. 9-2. Alternative hi-fi tone control circuit with separate high and low frequency feedback. Potentiometer Rl is the volume control Potentiometer R7 is the treble control and potentiometer RIO the bass control
Table 9-2. Parts List for Fig. 9-2.
100 KQ log pot 100 Q 18Q 180 a 27 a
Cl C2 C3
R6 R7 R8
R9 RIO Rll
C6 C7 C8 C9 C10 Cll C12
Rl R2 R3
10 KO log pot 150 a 10 Ka log pot 15 a
0.1 /iF 100 jtF 100 iiF 500 jiF 100 a»F 82 pF 1000/tF 0.1 ^F 0.15 pF 2pF lpF 2.2 /tF
Tone control network
Fig. 9-3. Simple hi-fi tone control circuit. Component values are determined for a sup ply voltage of 32 volts. Potentiometer R2isthe bass control. Potentiometer RSisthe treble control. Components within the dashed outline comprise the tone control network.
Hz and 10 kHz respectively in the extreme positions of the potentiometer. A (Baxandall) hi-fi tone control circuit associated with another
type of op amp is shown in Fig. 9-3. (Parts list is in Table 9-3.) The IC in this case is the CA3140 BiMOS op amp. The tone control drcuit is conventional and only a few additional discrete components are required to complete the amplifier circuit around the IC. This circuit Table 9-3. Parts list for Fig. 9-3. Rl
5 MO log pot 240 KQ
51 KQ 5 MQ linear pot
R6 R7 R8 R9
51 KQ 2.2 MQ 2.2 MQ 2.2 MQ
IC Cl C2
CA3140 750 pF 750 pF 20 pF 0.1 ftF
C3 C4 C5 0.1 itF Coupling Capadtor (C8) 0.047/tF
Fig. 9-4. Another hi-fi tone control circuit Potentiometer R4 is the treble control Potentiometer R6 is the bass control Supply voltage is 30 volts.
is capable of ± 15 decibels bass and treble boost and cut at 100 Hz and 10 kHz respectively. An alternative circuit using the same IC and giving a similar performance is shown in Fig. 9-4, with the parts list shown in Table 9-4. Both of these circuits require a supply voltage of 30 to 32 volts. Figure 9-5 shows the same circuit modified for dual supplies. Table 9-4. Parts list for Fig. 9-4.
Cl C2 C3 C4 C5 C6 C7 C8
0.1 jtF 0.01 jtF 100 pF 100 pF 0.001 /»F
0.002 ,tF 0.005 fiF
R6 R7 IC
5.1 mO 2.2 MO
18 KQ 200 KQ linear pot 10 KQ 1 MQ log pot 100 KQ CA3140 CA3140
IC—CA3140 Vin C1
Fig. 9-5. Tone control for dual supplies.
For the Shop
Constant Current or Constant Voltage Source A useful circuit employing the CA3018 integrated circuit array is shown in Fig. 10-1. This array comprises four transistors (two interconnected as a super-alpha pair) and four diodes. Tapping the super-alpha pair of transistors, a constant current source can be produced, the magnitude of this current being set by adjustment of the potentiometer Rl over a range of about 0.2 mA to 14 mA, de pending on the actual supply voltage. The same integrated circuit can also be used as a constant voltage source—Fig. 10-2. In this case, the constant voltage output is the zener voltage of the transistor that is worked as a zener diode, which is approximately 6 volts.
R1—10 Kft potentiometer
R3—3.3 Kft R4 is the resistance of the load through which the
constant current is to flow.
Fig. 20-2. Constant current circuit using components found in CA3018 array.
18 volts Output
Constant 6 volts I I
R1-3.3 KO R2-4.7 KO
Fig. 10-2. Circuit giving a constant 6 volts output from a 9- to 18-volt supply voltage, again using the CA3018 array.
Digital-to-Analog Converter The parts and equipment that you will need to perform this series of experiments are listed in Table 11-1. The oscilloscope is optional.
The first thing you need to do is devise some source of a digital signal to convert into an analog voltage. The simple circuit shown in Fig. 11-1 will create a 4-bit work dependent on the position of the switches. If a given bit's switch is in the upper position, it will tap off approximately 5 volts from the voltage divider. This represents a logic 1.
Moving the switch to the lower position grounds out that bit. The output will be 0 volts, or logic 0. You can set the four switches to create any 4-bit binary word from 0000 to 1111.
The LEDs and their associated resistors are optional. If a giv en LED is lit, that bit is at logic 1. A dark LED implies a logic 0 output for that bit. The resistors are used to limit the current through
Three basic resistor values are used in this circuit. The resistor marked RA should have a value of 33 K. The resistor marked 1^ should have a value of 18 K. The resistors marked ^ may have values ranging from 220 to 470 ohms. A smaller value will cause
the LED to glow more brightly. All ^ resistors should have the same value for a uniform display.
741 op amp IC 220 to 470 ohm resistor (see text) (need 4) 5.6 K resistor (need 4) 10 K resistor (need 2) 12 K resistor (need 6) 15 18 22 33 39 82
K K K K K K
resistor resistor resistor resistor resistor resistor
Table 11-1. Parts list for Project 11.
LED (need 4) SPDT switch (need 4) hook-up wire voltmeter
Fig. 12-1. This simple circuit can simulate 4-bit digital words for use in Project 11.
Breadboard the digital output simulator of Fig. 11-1 and the D/A converter circuit shown in Fig. 11-2. Use the following component values:
Rl = 82 K resistor R2 - 39 K resistor R3 = 22 K resistor 56
R4 = 10 K resistor R5 = 10 K resistor
R6 = 15 K resistor
Fig. 11-2. This D/A converter circuit is explored in the first part of Project 11.
Bit A is the least significant bit (2°, or 1). Bit D is the most significant bit (23, or 8). The 4-bit binary words will be written in the following order: DCBA
- (1 x 23) + (0 x 22) + (1 x 21) + (1 x 2°) = (1 x 8) + (0 x 4) + (1 x 2) + (1 x 1) =8+0+2+1 = 11
Any input bit at logic 0 will be feeding 0 volts into the input of the op amp. That bit will have no effect on the output. Set the binary switches for an input of 0000. The output should be zero volts (or very close to it). Now raise the switch for input bit A, leaving the other three
switches grounded. This makes the input word 0001. The D/A converter functions as an inverting amplifier with its gain set by the ratio of R5 and Rl. R5
The negative sign simply indicates that the output polarity is inverted. Because you will be using only positive input voltages in this application, the output always is negative. The exact input voltage for a logic 1 input bit is set by the power supply voltage and the values of RA and RB in Fig. 11-1. Assuming the supply voltage is ±15 volts, and RA and RB have the values specified earlier (33 K and 18 K respectively), this works out to 57
approximately +5.3 volts. Component tolerances and variations in the supply voltage may cause slightly different values. For an input of 0001, the output voltage should be equal to the input voltage (5.3 volts) times the gain (-0.12). Vo = 5.3 x -0.12 = -0.636
Measure the output. Do you get a value close to this? Return bit A to logic 0, and change bit B to logic 1 (0010). The input resistor for this bit is R2. Cj =
Ideally this should be twice the gain of bit A (2 x -0.12 = -0.24), but because you are probably using standard resistor values, you can be satisfied with coming close. Component tolerances cause deviation from the nominal values in any event. Once again, a logic 1 input bit equals 5.3 volts, so the output voltage for an input of 0010 is equal to ViG. Vo = 5.3 x -0.26 - -1.36 volts
Does the voltmeter read an output voltage close to this calculated value? Ground bit B and raise bit C to logic 1 (5.3 volts). You should now have 0100 as your input binary work. The gain is set by R5 andR3.
G= -— = - 10'000 = .045 R3
Vo = 5.3 x -0.45 = -2.41 volts
Next, set the input word to 1000. All switches are grounded (logic 0) except for D. What output voltage do you get now? It should work out according to the following equations:
v« = G x D 58
= -1 x 5.3 - -5.3 volts
What happens if more than one input bit is at logic 1? In this case, the circuit acts like a summing amplifier. The inputs are weighted by the different value input resistors. The output of this circuit is equal to the following relationship:
-R5\ /,. -R5\ / -R5\ /.. -R5\ l+lViuX 1+1 VirX l+IVjaX I
R3 / \ -
For the component values you have been working with, the equation simplifies as shown here.
Vo = (^x -0.12) O^x -0.26)+ (Vfc x -0.45) + (Vw x-1) Set the input binary word to 0011. That is, A and B are each at logic 1 (+5.3 volts) and C and D are at logic 0 (0 volts). The output in this case should be simply the sum of A and B calculated separately. Plug the appropriate voltage values into the formula, and find the nominal output voltage. Vo = (5.3 x -0.12) + (5.3 x -0.26) + (0 x -0.45) + (0 x -1) - -0.636 + -1.36 + 0 + 0 - -1.996
For a 4-bit binary word, there are 16 possible combinations. Try the input to each combination and jot down your results in Table 11-2. Compare your results with the calculated values listed in Table 11-3. For the next part of this experiment, you will use the R-2R ladder D/A converter circuit illustrated in Fig. 11-3. Again the input is ob tained from the digital simulator circuit of Fig. 11-1. Use the following ies: component values: Rl R2 R3
R4 R5 R6
12K 22 K 5.6 K 12K 5.6 K
12K 5.6 K R9 12 K R10 12K Rll 5.6 K
The most significant bit (D = 23 = 8) passes through Rl and
R2 before being fed into the inverting input of the op amp. The gain 59
Table 11-2. Worksheet for Project 11. Input Word
0 o o
for this bit depends on the ratio of R3 and R2 and the input voltage that passes through a voltage divider made up of Rl, R4, R6, R8, and RIO. The input voltage is tapped off at point E. The input signal passes through Rl and then is split off between R2 and ground through R4, R6, R8, and RIO. For a logic 0 input, the voltage at point E will simply be 0. For a logic 1 input, a 5.3-volt signal is fed through the voltage divider. The total resistance to ground is a sum of the resistances. Rt
Rl + R4 + R6 + R8 + RIO
12,000 + 5600 + 5600 + 5600 + 12,000
According to Ohm's Law, the current flow should be equal to 0.13 mA. Table 11-3. Calculated Values for Project 11.
0 0 1
0 0 0 0
0 0 0 0 0 0 0 0
0 0 0 0
0 0 1
-2.00 -2.41 -3.05 -3.76 -4.41
-5.29 -5.94 -6.65 -7.30 -7.70 -8.35 -9.06 -9.70
Fig. 22-3. The second halfof Project 11 examines the operation ofa R-2R ladder type D/A converter.
= 0.0013 amp = 0.13 mA
The voltage drop across Rl equals 1.56 volts. Vx = I • Rl = 0.00013 x 12,000 - 1.56 volts
Subtract this value from the original input voltage (5.3 volts.) Ve - V, - Vx x 5.3 - 1.56 = 3.74
This is fed into the inverting amplifier. The gain for the D input is calculated as follows: -R3
= -1.8333 = -1.8
The negative sign simply indicates that the output polarity is reversed because the op amp's inverting input is being used. The output voltage when input D is at logic 1 and the other three
inputs are at logic 0 (1000) should be close to -6.86 volts.
Vo » VB x GD - 3.74 x -1.8 - -6.86 volts
The situation is slightly more complex for the other input bits, but you can simplify matters by looking at each one individually. Fig ure 11-4 simplifies the schematic for input C. Inputs A,B, and D and their related components are ignored.
The input signal is fed through a voltage divider made up of R5, R6, R8, and RIO. The input voltage is tapped off at point F between R5 and R6. The input resistance to the inverting amplifier is the se ries combination of R4 and R2.
Solve for the input voltage by first finding the current with Ohm's
(R5 + R6 + R8 + RIO) 5.3
(12,000 + 5,600 + 5,600 + 12,000) 5.3
0.00015 amp = 0.15 mA
VF = V, - Vc = Vj -IRS - 5.3 -(0.00015 x 12,000) « 5.3 -1.8 = 3.5 volts
Fig. 11-4. Input C sees this equivalent circuit.
Next, find the gain for input C. r
So, the output voltage for C = 1 with the other bits = 0 (0100) works out to about -4.375 volts.
VQ = VpxGc = 3.5 x -1.25 - -4.375 volts
Figure 11-5 shows the equivalent circuit for input B. Running through the equations for this input bit you find the following: VV
(R7 + R8 + R10) 5.3
(12,000 + 5,600 + 12,000)
= V, -V, - V, - IR7 - 5.3 -(0.00018 x 12,000) = 5.3 -2.15 = 3.15
Fig. 11-5. This equivalent circuit is seen by input bit B.
(12,000 + 5,600 + 5,600)
Vo - VG x GB = 3.15 x -09.5 = 2.99 volts
Finally, solve the equations for bit A. See Fig. 11-6. V,
Vh » V, - VA
(12,000 + 12,000)
= 0.22 mA
= V, - IR9 - 5.3 - (0.00022 x 12,000)
5.3 - 2.65 = 2.65 volts R3 (R2+R4+R6+R8) 22,000
(12,000 + 5600 + 5600 + 5600)
T Fig. 11-6. Here is the equivalent circuit for input bit A.
vo " vh * ga = 2-65 x -°-76 = -2-02 volts Measure the output voltage for several combinations of digital inputs. There will be some imbalance of the steps because 12 K is not exactly twice 5.6 K, as required by the R-2R resistor network. For practical applications, low tolerance precision resistors should be used.
Logic Probe This project is a piece of test equipment that can be used to study and (if necessary) troubleshoot most of the projects presented in this book, and other digital circuits too. The project is called a logic probe. Logic probes are simple but powerful tools for analyzing what goes on in a digital electronics circuit. Many commercially available However, this inexpensive do it-yourself logic probe will come in handy for any digital elector
in special features, it more than makes up for with low cost. The project should not cost you more than two or three dollars. A logic probe is simply a device that allows you to determine the logic state (0 or 1) at any point in a digital circuit. A super-simple two-component logic probe is shown in Fig. 12-1. The ground lead can be fitted with an alligator clip so that it can be connected to the same ground as that of the circuit being tested. The probe is a short length of stiff solid wire, or use a common test lead probe. If this probe is touched to a point in the circuit with a logic 1 (high voltage) signal, the LED lights up. At all other times, the LED remains dark.
The resistor is used for current limiting. If the LED is allowed to draw too much current, it can be damaged. This resistor generally has a value of less than (1000 ohms), with 330 ohms being a typical value. This super-simple logic probe can be used without modification for any of the major logic families (CMOS, TTL or any of its
Fig. 12-1. A super simple logic probe can be made from a resistor and an LED.
variations, DTL, etc.). Because the circuit contains no active logic elements and takes its power from the circuit being tested, it is universal. This circuit can be whipped together for under a dollar. While
functional, this simple approach leaves a lot to be desired. For one thing, it does not give a definite indication of a logic 0 state. If the LED does not light, you could have a logic 0 signal, or the probe might not be making good contact with the pin being tested. There might even be a broken lead, or the LED itself could be damaged. With the circuit of Fig. 12-1, there is no way of telling. Another potential problem area with this circuit is the possibility of excessively loading the IC output being tested. This can especially occur when the gate in question is already driving dose to its maximum fan-out potential. Both of these problems can be side stepped by adding a pair of inverters and a second LED/resistor combination, as illustrated in Fig. 12-2. When a logic 0 is applied to the probe tip, the first inverter changes it to a logic 1, lighting LED A. The second inverter changes the signal back to a logic 0, so LED B remains dark. Conversely, when a logic 1 is fed into the probe tip, the output of the first inverter is a logic 0, so LED A stays dark, and LED B is lit up by the logic 1 signal appearing at the output of the second inverter. This circuit can give a definite indication of either a logic 1 or a logic 0 signal. If neither LED lights up, the probe is not making proper contact. This circuit is much less ambiguous than the earlier
But that's not all this improved logic probe can tell you. This device can also indicate the presence of a pulsating signal (keeps reversing states). If the LEDs alternately blink on and off, a low frequency pulse is being fed to the probe. If both LEDs appear to be continuously lit, a high frequency pulse signal is indicated. 67
Actually, in this case the LEDs are still alternately blinking on and off, but it is happening far too fast for the eye to see, so the LEDs look like they are both staying on at all times. The circuit of Fig. 12-2 calls for just five components—two LEDS, two resistors (330 to 1000 ohms—the value is not critical), and a hex inverter IC. Two of the six inverter sections are used in this circuit. The other four inputs should be grounded to ensure circuit stability. This is particularly important if a CMOS chip is being used. Like the simpler version of the logic probe discussed earlier, this circuit steals its power from the circuit being tested. Alligator clips on the power supply leads can be attached to the power supply output of the circuit being tested. A CMOS CD4009A hex inverter IC is the best choice, because it can be driven by either TTL or CMOS devices. By tapping into the tested circuit's power supply, the voltage levels will be automatically matched. In some cases, it might be desirable to add a pull-up resistor at the probe's input, as shown in Fig. 12-3. The value of this resistor is not critical, but it should probably be kept in the neighborhood of 1 K (1000 ohms). If you intend to work with just TTL ICs, you can substitute a 7404 hex inverter (or the equivalent in the appropriate subfamily, like the low-power Schottky 74LS04). Use the pin numbers that are
To ground of power supply
Fig. 22-2. An improved logic probe uses inverters to prevent circuit loading and to display both logic 1 and bgic 0 signals.
Fig. 22-3. If a pull-up resistor is included in your logic probe, a CMOS probe can be used on TTL circuits.
shown in parentheses in Fig. 12-2. This pin marked (*) has no equivalent on the 7404 and should simply be ignored. I strongly recommend that you build a permanent version of this project. It will be useful throughout your project-building endeavors.
Digital Capacitance Meter
digital ICs. The process is not all that different from the digital voltmeters described in the previous chapter. A basic digital capaci tance meter circuit is shown in block diagram form in Fig. 13-1. The first stage is a monostable multivibrator. You should recall that a monostable multivibrator has one stable state. Assume the stable state is logic 0. The output of the monostable multivibrator remains at logic 0 until the circuit is triggered. At that time, the output switches to logic 1 for a specific period of time that is defined by a resistor/ca pacitor combination. After this period of time, the output of the multivibrator returns to its stable state (logic 0). In this application, the timing resistor is a fixed value. (In some cases, different resistors may be switch-selectable for different ranges.) The timing capacitor is the unknown capacitance being measured. Therefore, the output of the monostable multivibrator goes to logic 1 when triggered for a period of time that is directly proportional to the input capacitance. The rest of the circuit is very similar to the digital voltmeters described in the last chapter. The output of the monostable multivibrator controls an AND gate that blocks or passes the reference oscillator signal through to the counter stage. The count is checked for over-range, decoded, and displayed. 70
Reset R ti
Fig. 13-1. A digital capacitance meter is quite similar to a digital voltmeter (Project 16).
Because the monostable multivibrator's on time is directly proportional to the unknown input capacitance, the count displayed will also be proportional to the value of Cx. A pushbutton to reset the counter and manually trigger the monostable multivibrator is usually mounted on the front panel of the instrument. A practical capacitance meter circuit is illustrated in Fig. 13-4. The parts list is given in Table 13-1. Because the voltage across the test points (and therefore through) Cx is less than two volts, the measurement process is safe for virtually any component. This unit is capable of measuring capacitances from less than 100 pF to well over 1000 /*F. If electrolytic capacitors are to be tested, be sure to hook up the meter with the correct polarity. Point A should be attached to the capacitor's positive lead, and point B is negative. Resistor Rl sets the full scale reading for the meter. This component should be a trim pot that is set during calibration, then left alone. Once the necessary resistance is found, the trim pot could be replaced with an appropriate fixed resistor to eliminate the need for periodic recalibration. Smaller resistance values should be used to measure larger capacitances. A100 K pot could be used for a IX scale, while a 10X scale would be better served with a 5 K potentiometer. Multiple resistances could be set up for switchselectable ranges.
USES FOR A CAPACITANCE METER
The obvious use for a digital capacitance meter is to determine the value of unmarked capacitors or to check to make sure a capad71
Fig. 13-2. The schematic for a complete digital capacitance meter.
11 8 9 12 14
6 2 17
6 2 17 13121110 9 1514
a b c d efg
Table 13-1. The Digital Capacitance Meter Project Calls for These Components. 555 timer CD4011 quad NAND gate
IC1 IC2, IC8 IC3, IC4, IC6 IC5, IC7
74C90 decade counter
CD4511 BCD to 7-segment decoder NPN transistor
1N4734 diode (or similar) LED (overflow indicator)
D1 D2 DIS1.DIS2 R1 R2.R5 R3, R4, R7 R6. R8 R9, R10 R11 -R24 C1.C2 C3
common cathode seven segment LED display calibration trimpot—see text
2.7 k Va watt resistor 15 k 1/4 watt resistor
10 k1/4 watt resistor 1.8 k Va watt resistor 330 ohm Va watt resistor 0.047 yJF capacitor 0.1 /xF capacitor 0.01 fxF capacitor 0.0022 /xF capacitor DPDT normally open pushbutton
C4.C5 C6 S1
—push to clear and test
tor is true to its marked value. The capacitor's tolerance can be calculated with the following formula:
-—1 x 100
where T is the tolerance (or percent of error) in percentage, CN is the nominal, or marked value for the capacitor, and CM is the ca pacitance value measured.
As an example, let's assume we have a capacitor that is marked 5 pF. When we hook this component up to our digital capacitance meter, we get a reading of 4.357 /«F. The tolerance is therefore equal
(ABX(CM-CN)/C,,) x 100
= = = 74
(ABS(4.357 -5)15) x 100 (ABS(-0.643/5) x 100 (0.643/5) x 100 0.1286 x 100 12.86%
This is not bad for an electrolytic capacitor many of which often have rather wide tolerances, but it obviously would be a poor choice for any application requiring precision. A digital capacitance meter can also be put to use in a number of other interesting ways. For instance, this type of instrument can come in handy for tracking down stray capacitances that can cause problems in many circuits, especially those operating at high frequencies. Printed circuit boards with closely placed traces are of ten subject to stray capacitance problems. Components other than capacitors often exhibit internal capacitances that may need to be taken into account when designing precision circuits. For example, in an rf amplifier circuit, the transistor base-to-collector and emitter-to-collector capacitances can cause instability and/or oscillation in some cases. Cables that consist of more than a single conductor have a nat ural capacitance per foot. These cables include coaxial cable, antenna twinlead, and ribbon cables. Because a capacitor is basically two conductors separated by an insulator, multi-line cables naturally behave as long capacitors. The capacitance per foot of a cable can be determined by measuring a known length of the cable with a capacitance meter, and using the following formula:
where Cf is the cable's capacitance per foot, C is the measured ca pacitance, and F is the number of feet in the measured sample. Cf and C will always be in the same units. If C is measured in picofarads, Cf will also be in picofarads. Or, if C is in microfarads, Cf will be in microfarads too.
Let's say we have a 2 Vi-foot sample of a cable. When you measure the capacitance of this length, you get a reading of 55 pF. Now, you can easily calculate the capacitance per foot. Cf = C/F = 55/2.5 = 22 pF per foot. By rearranging this formula, you can determine the total capac itance of a length of cable if you know the capacitance per foot, and the length: CT = F x CF 75
where CT is the total capacitance, F is the length in feet, and CF is the capacitance per foot. As an example, assume you have a 235-foot length of the cable with the same capacitance as in the last example (22 pF per foot). The total capacitance works out to F x CF - 235 x 22 - 5170 pF. Another algebraic manipulation of this same formula allows us to determine how long an unknown length of cable is. This could
be necessary if the cable is buried, embedded in a wall, or otherwise inaccessible for direct measurement. To perform the calculation, you need to know the capacitance per foot, and then take a reading of the cable's total capacitance. The formula is:
Let's say you need to determine an unknown length of a piece of the 22 pF per foot cable. Measure the total capacitance of the cable to get a reading of 4756 pF. The length of the cable is therefore equal to C^Cp = 4756/22-just over 216 feet, 2 inches. Another novel application for a digital capacitance meter is as a digital thermometer. For this application, a high quality capacitor with a known temperature coefficient is needed. This specification is usually expressed as x parts per million per degree centigrade. For purposes of illustration, assume you have a capacitor with a wide temperature coefficient of 100 parts per million per degree centigrade. K this capacitor measures 0.1475 fiF at 10 degrees centigrade, at 20 degrees centigrade it should produce a reading of 0.1465 fiF. As you can see, a digital capacitance meter can be an extremely handy instrument to have on your workbench.
Digital Frequency Meter Most of the commercially available frequency counters around today use the "window" counting method. A sample of the input signal is allowed through a gate. This sample lasts a specific and fixed period of time. By counting the pulses during this sample period, the input frequency can be determined.
A block diagram for a "window" type digital frequency meter is shown in Fig. 14-1. The input signal is first fed through an amplifier stage to boost the signal to a usable level. An amplifier stage is not always used, but its presence improves the sensitivity of the instrument, allowing lower signal levels to be measured accurately. The next stage of the circuit is a Schmitt trigger to convert any input waveshape to a rectangle wave that can be reliably recognized by the digital circuitry. If only square or rectangle waves are to be measured, the Schmitt trigger stage may be omitted. This processed input signal is fed to one input of an AND gate• The other input to the gating circuit comes from a reference oscillator, or timebase (as it is usually called in this application). The timebase feeds out three signals (or a single signal tapped off with delay circuits, as shown in the diagram). These three timebase signals are synchronized, and their timing relationship are critical. The three signals are illustrated in Fig. 14-2. The first signal (labeled GATE) is fed to the input of the gating
circuit, effectively opening (when logic 1) and closing (when logic
Gate \ Latch Reset
Decoders Delay 2 Displays
Fig. 24-2. The bask digital frequency meter is another variation on the basic digital voltmeter.
0) the "window/' allowing the input pulses to be counted. The sec ond signal, which is delayed until after the first is over, latches the output of the counters so they can hold their final value long enough to produce a readable display, while the third signal resets the counters to zero for the next measurement cycle. If the output latch ing was not done, the counter would count up to the appropriate amount, then immediately jump back to 000 and start over, never producing a stationary reading. With the latches, only the desired final count from each measured cycle is displayed.
Incidentally, the accuracy of most digital frequency counters is given as x% ± 1 digit. The least significant digit might bob up and down on successive measurement cycles. This happens because a partial input pulse can get through the "window," as illustrated in Fig. 14-3.
The timebase oscillator must be very precise in its output frequency with as little frequency drift as possible. Crystal oscillators are often used. The input frequency being measured must be higher than the reference frequency. If the input frequency is lower than the reference frequency, only 1 or 0 pulses can get through each 78
Fig. 14-2. Staggered pulses are required from the timebase oscillator in a digital frequency
4 'window,'' which obviously would not result in a meaningful reading. To measure lower frequencies, an additional frequency multiplier input stage can be added between the Schmitt trigger and the gate. Similarly, to measure very high frequencies that would over-range the counter stages, a frequency divider stage could be added to drop
the input signal to a lower frequency. Input
innnr u u u u Window 1
Count = 5
LIU LJ U Window 2
Count = 6
I i i
Fig. 14-3. Partial pulses appearing during the "window" counting time may cause
some bobbing of the least significant digit.
Fig. 14-4. A complete digital frequency meter is shown in this schematic.
Most commercial frequency counters have three to six counter stages for maximum counts of 999 to 999999. Switchable frequency multipliers and/or dividers are also generally included to allow manually selectable ranges. Decimal points may or may not be included in the display readout.
A DIGITAL FREQUENCY METER CIRCUIT
A complete digital frequency meter project is illustrated in Fig. 14-4. Table 14-2 is the parts list for this circuit. Each digit of the 80
display is driven by a CD4026 decade counter (IC2 through IC5). Four digits are indicated in the diagram, but it is a simple matter to extend the display to contain additional digits. Simply connect pin 5 from the last stage to pin 1 of the following stage. The other pins of each IC are connected in the same way as for ICs 2 through 5. Ql and IC1 pre-condition the input signal so that it will have an acceptable level and waveshape to be reliably counted by the digital circuits. IC6 is wired as a reference oscillator, whose output frequency can be adjusted with R33, a 1MO trim pot. Calibration is done by applying a known frequency source to the input of the circuit and adjusting R33 for the correct reading. 81
Table 14-1. Parts list lor Digital Frequency Meter.
4583 Schmitt trigger
CO4026 decade counter 556 dual timer CD4011 quad NAND gate
R1 R2 R3, R39 R4 R5-F&2 R33 R34 R35-R38
NPN transistor (2N3302,2N5826, Motorola HEP-728, Radio Shack RS-2013. or similar)
22* resistor 18 k resistor 100 k resistor 10 megohm resistor 220 ohm resistor 1 megohm trimpot 470 k resistor 10 k resistor 1 fif 35 volt electrolytic capacitor 10 iiF 35 volt electrolytic capacitor 0.001 mF disc capacitor
Multiple-Output Power Supply This project can be used to power many of the experiments presented throughout this book. You can also use it to power other projects of your own. The circuit is shown in Fig. 15-1 and a parts fist is given in Table 15-1. Really, there isn't much to say about this circuit. There are 6 voltage outputs (with respect to ground): +12 volts regulated +9 volts +5 volts regulated
-5 volts -9 volts -12 volts regulated
Notice that three of the outputs are directly regulated and should be almost exactly at their nominal values. The other three outputs are derived via resistive voltage dividers. Their values might be slightly off. Measure each of the outputs with a voltmeter. You might also want to check for output ripple with an oscilloscope. Without heatsinking, each regulator should be able to supply about 0.5 amp without shutting down. Bear in mind that several outputs are taken off some of the regulators. The total current draw
should not exceed 0.5 amp for each regulator. That is:
Km + I+9V = 0.5 amp
I+5V = 0.5 amp
I-5V + I-9V + I-12V = 0.5 amp 83
Fig. 25-2. Multiple output power supply circuit.
Table 15-1. Parts List for Multiple-Output Power Supply. Voltmeter Oscilloscooe (ODtional) 1
1 1 1 1
D1-D4 C1,C8 C2, C3, C4, C5, C9, C10 C6.C7.C11 R1.R3 R2 R4 R5
ac line cord 36-volt, 3 amp transformer—center tapped 3-amp fuse & holder 7805 +5 volt regulator IC 7812 +12 volt regulator IC 7912 -12 volt regulator IC 1N4001 diode 500 mF 50 volt electrolytic capacitor 0.1 pF capacitor 10 pF 25-volt electrolytic capacitor 2.2 K resistor (2200 ohms) 6.8 K resistor (6800 ohms) 2.7 K resistor (2700 ohms) 3.9 K resistor (3900 ohms)
You can get greater output current by adding heatsinks to the regulators (you will need a heavier duty transformer and larger fuse too). Don't try to get more than one amp out of each regulator IC.
Dc Voltmeter In this experiment, you use an op amp to make a dc voltmeter. The parts and equipment you need are listed in Table 16-1. Start by breadboarding the circuit shown in Fig. 16-1. Resistors Rl through R7 form a voltage divider network to provide a number of voltages to measure during the experiment. Use the following resistor values in the voltage divider: Rl R2
33K 2.2 K 10K IK
R5 R6 R7
22 K 10K 33K
The op amp is connected as a simple unity-gain non-inverting voltage follower with a voltmeter connected to the output. Connect the op amp's non-inverting input (V) to each of the points in the voltage divider network (labeled A through F). Write each measured voltage in the appropriate space in Table 16-2. Compare your results to the calculated values listed in Table 16-2. Your measured voltages should be dose to the ones listed here. There might be some deviation from the nominal values due to component tolerances op amp offsets, and minor measurement errors.
It's a simple matter to add gain to an op amp dc voltmeter sim ply by adding an input and a feedback resistor, as illustrated in Fig. 86
Table 16-1. Parts list for a Dc Voltmeter.
Breadboard system 741 op amp IC 1 K resistor (need 2) 2.2 K resistor 3.3 K resistor 4.7 K resistor 10 K resistor (need 4) 22 K resistor (need 2) 33 K resistor (need 2) 100 K resistor (need 3) voltmeter (15 volt range) hook-up wire
Fig. 16-1. This voltage divider network and unity gain dc voltmeter are used for the first part of Project 16,
Measured Output Voltage
Table 16-2. Worksheet for the First Part of Project 16.
C D E F
Fig. 16-2. This project also demonstrates that a dc amplifier can have gain.
Table 16-3. Calculated Values for the First Part of Project 16.
A B C D E F
Calculated Output Voltage +6.10 +5.50 +2.81 +2.54 -3.40 -6.10
volts volts volts volts volts volts
16-2. Remember to turn off the power supply while making changes to the circuit. Use a 10 K resistor for R8 and a 4.7 K resistor for R9. Since this is a non-inverting amplifier circuit, the gain is as follows:
1 + 0.47
Repeat the first part of the experiment, recording your values in the appropriate column in Table 16-4. Now change R9 to a 10 K resistor. Leave R8 alone. This makes the gain equal to Z. 10,000
Table 16-4. Calculated Values for the Second Part of Project 16. Input Point
(R9 = 4.7 K)
(R9 = 10 K)
(R9 = 22 K)
A B c D E F
Repeat the first part of the experiment, recording your values in the appropriate column in Table 16-4. Now change R9 to a 10 K resistor. Leave R8 alone. This makes the gain equal to Z. 2.2 = 3.2
Repeat the measurements and record your results in the table. You might have found some of the measurements forced the op amp into saturation. This is to be expected. You must be careful not to exceed the op amp's maximum output voltage in using a practical dc voltmeter. Compare your results with the calculated values listed in Table 16-5. A dc amplifier with gain is obviously most useful when the volt
age to be measured is quite small. Small voltages may be difficult to measure directly. Return to the circuit shown in Fig. 16-1. This time, use the following resistor values in the voltage divider: Rl
2.2 K 3.3 K
100 K 100 K
Try measuring the tap-off points on the voltage divider using the unity gain dc voltmeter of Fig. 16-1. The analog voltmeter at
the output should have a measurement range of about 15 volts. Table 16-5. Worksheet for the Third Part of Project 16. Input Point
(R9 = 4.7 K) A B C D E F
+8.97 +8.09 +4.13 +3.73 -5.00 -8.97
volts volts volts volts volts volts
(R9 = 10K)
(R9 = 22 K)
+12.20 volts (*) +11.00 volts +5.62 volts +5.08 volts -6.80 volts -12.20 volts*
+19.52 (*) +17.60 (•) +8.99 volts
* The op amp may be driven into saturation, causing clipping of the output voltage.
+8.13 volts -10.88 volts -19.52*
Table 16-6. Measured Values for the Third Part of Project 16. Measured Output Voltages
Gain = 11
B C 0 E F
Table 16-7. Calculated Values for the Third Part of Project 16. Measured Output Voltages Input Point A B
C D E F
Unity Gain +0.86 +0.72 +0.41 -0.06 -0.20 -0.86
volt volt volts volt volt volt
Gain = 11
+9.46 volts +7.92 volts +4.51 volts -0.62 volt -2.18 volts -9.46 volts
Record your results in Table 16-6. You will probably have some difficulty reading the meter. The differences will be slight, and the meter's pointer won't move very far. Now use the dc voltmeter with gain shown in Fig. 16-2. Use a 10 K resistor for R8 and a 100 K resistor for R9. This gives the following gain:
G = 1 +
-1 + 10-11
Measure the voltage at each of the tap-off points on the voltage divider and compare your results in Table 16-6. These voltages are easier to read, aren't they? Compare your measured voltages with the calculated values shown in Table 16-7. In a practical dc voltmeter, you would draw a new scale to place on the meter face so that you could read the input voltage directly, even though it is being multiplied by the gain of the op amp. 91
Index A alarm, car thief, 3 AM/FM radio, 20-25 amplifiers audio, 26-31 power, 32-41
audio amplifier, 26-31 LM380 IC circuit in, 33, 38 modifications to, 28
B Baxandall hi-fi tone control circuit,
47 bridge configurations, 40 LM380 in, 41
cables, determining capacitance of, 75 capacitance meter, digital, 70 car thief alarm, 3 cascading, 26 dock, 9-19
constant current, 53 constant voltage source, 53 converters, digftal-to-analog, 55-65 corner frequency, 40
dc voltmeter, 86-91 digital capacitance meter, 70-76 uses for, 71 digital clocks, 9-19 digital frequency meter, 77-82 digital voltmeter, 71 digital-to-analog converter, 55-65
E electric motor speed controller, 5 frequency meter, digital, 77-82
hi-fi tone controls, 44-49 home projects, 1-49
AM/FM radio, 20-25
audio amplifiers, 26-31 car thief alarm, 3 clock, 9-19
electric motor speed controller, 5
hi-fi tone controls, 44-49 intercom, 1 power amplifiers, 32-41 tape player amplifier, 42 intercom, 1
logic probe, 66 loudness control, 44
multiple-output power supply, 83
O Ohm's law, 60, 62 power amplifiers, 32-41 power supply, multiple-output, 83 projects AM/FM radio, 20-25 audio amplifiers, 26-31 car thief alarm, 3 clock, 9-19 constant current or constant
voltage source, 53 dc voltmeter, 86-91 digital capacitance meter, 70-76 digital frequency meter, 77-82 digital-to-analog converter, 55-65
electric motor speed controller, 5 hi-fi tone controls, 44-49
intercom, 1 logic probe, 66 multiple-output power supply, 83 power amplifiers, 32-41
tape player amplifier, 42
Q quartz crystal oscillator, 9 R radio, AM/FM, 20-25 RIAA equalization, 39
S Schmitt trigger, 77 shop projects, 51-91
constant current or constant
voltage source, 53 dc voltmeter, 86-91 digital capacitance meter, 70-76 digital frequency meter, 77-82 digital-to-analog converter, 55-65
logic probe, 66
speed controller, electric motor, 5
T tape player amplifier, 42 timebase frequency, 9 timebase signals, 78 timers, 9
tone controls, 44-49
V voltage gain, 31 voltmeter dc, 86-91 digital, 71
volume control, 44 W window counting method, 77, 79
Selected Index 14583 Schmitt trigger, 82 1N4734 diode, 74 2N3302, 82
2N5826, 82 555 timer, 74 556 dual timer, 82 74C90 decade counter, 74 74LS04 low-power Schottky, 68 CA3018IC, 53 CA3035 IC, 26 CA3140 BiMOS op amp, 47 CD4009A hex inverter, 68 CD4011 quad NAND gate, 13,15,74, 82
CD4017 decade counter, 12,13,15
CD4026 decade counter, 81, 82 CD4049 hex inverter, 15 CD4081 quad AND gate, 13 CD4511 BCD to 7-segment decoder, 15,74 CD4518 dual BCD counter, 15 LM380 audio amplifier IC, 33 MM5369 timer IC, 10 TCA600/900, 5 TCA610/910, 5 TCA8305 IC, 44 TCA830S op amp, 1 TDA1071, 20 TDA1151 IC, 5 TDB0556A dual timer, 3
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DeltonT.Horrfs All-Time Favorite Electronic Projects Provides many hours of fun, challenging, hands-on electronics experience! This is a collection of some of Horn's most fascinating electronic projects. Avoiding overly theoretical explanations, he provides stepby-step instructions, easy-to-follow drawings, diagrams, and sche matics for each project. The list of projects includes: • Intercom
• Car Thief Alarm
Electric Motor Speed Controller
Digital Frequency Meter
• AM/FM Radio
• Power Amplifiers
• Audio Amplifiers
• Hi-Fi Tone Controls
• Tape Player Amplifier •
Delton T. Horn is an experienced electronics hobbyist and tech nician. He is the author of numerous popular electronics and com puter titles including Designing IC Circuits . . . With Experiments, Amplifiers Simplified, and 101 Solderless Breadboarding Projects.
$7 . TS
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